I have used a simple dipole for 14MHz temporarily. This antenna was not much as quiet as the Deltaloop but much simpler to install and fix, compared to the loop that has been my standard antenna for 14MHz for years. The noise in the receiver when using the dipole was beyond words.
Thus I changed the dipole for a quad loop again the last weekend. When connecting the transceiver I found that in spite of good SWR (1:1.2) the transmitter was self-oscillating. With the spectrum analyzer I could recognize that there were oscillations in the medium wave range as well as around 14 MHz. This made me examining the transmitter more closely.
After estimated a dozen of checks and failed attempts I changed to improve the performance I decided to alter the input transformer of the final stage. This item, which by origin had a winding ratio of 4 to 2+2, then was changed by a transformer having 3 to 3+3. This even deteriorated the situation.
Next I concluded that the second winding might have too high impedance. So I went to a 4 to 1+1 transformer (pig nose core BN43-202).
This finally cured the problem to a 100% and gave a very stable transmitter. Output power increased to 28 watts when fully modulated. This might be the direct outcome of better impedance matching.
Here is the current transmitter circuit with all the improvements made so far:
The first power amplifier for this transceiver project initially was capable to produce 20 watts of SSB pep rf power on the 20 meter band. The transistors in use are the 2SC1969 bipolar types by eleflow.com. A single device is rated to max. output power of 16 watts according to data sheet. In push-pull mode this nearly doubles because each of the semiconductors only has to amplify half of the duty cycle. Hence I started trying to get a little bit more power out of the PA assembly.
First I modified T3. The former data was: 1+1 turn primary center tapped and 4 turns secondary on a homemade pig nose core of 2×3 stacked FT50-43 toroids.
The new transformer has got 1 secondary turn more. Wire is 0.8mm enam.wire both for primary and secondary.
First I tried out 0.5mm enam. wire for the secondary which resulted in about 23 watts output pep. Next I used 0.8 mm diameter enam. wire which reduces the consequences of skin effect significantly . Penetration depth of a 14MHz ac current is only some 50 µm, thus every increase in surface area reduces resistance.
This set of measurements greatly improves two figures:
Output power increases from 20 to 25 watts pep,
carrier suppression improves by 5 dB.
The enhanced signal on the RIGOL DS1054 scope:
About 100 volts pp. (i. e. 35.5 volts rms) at 50Ω equals to about 25 watts pep. Nice improvement for a minor change that was done within 20 minutes!
A general and good practice in engineering is a steady process of improvement. This article describes the construction of a high performance transmitter/receiver for SSB (voice) communication covering the 14MHz (20 meters) high frequency amateur radio band.
Various modules that have proven high performance, liability and ruggedness in recent constructions will be combined to form a radio with outstanding receiver performance, an ultra linear transmitter with output range 15 to 20 watts and a top audio sound quality both on transmit and receive.
Key features are:
Dual DDS frequency generation with AD9834 (Local oscillator) and AD9951 (VFO),
Microcontroller (MCU): ATMega644P by ATMEL,
Single conversion superhet receiver with 9MHz interfrequency (IF) and preamplifier, mixer and IF amplifier equipped with Dual-Gate-MOSFETs,
Audio-derived automatic gain control (AGC),
Transmitter with MC1496 as double sideband (DSB) modulator and NE602 as transmit mixer,
power transimitter with 4 stages, final stage in push-pull mode.
Another version of this radio has been built before. But this one was equipped with a variable frequency oscillator (VFO) because of nostalgia reasons. Unfortunately a VFO lacks certain features (frequency stability above all) which can be overcome by using digital frequency synthesis without losing performance. Usage of a high performance DDS systems is a prerequisite to achievement and a possible solution.
Most building blocks of that respective radio have been redesigned except the VFO section that turned out as not being able to deliver the projected frequency stability to a 100% degree. Frequency instability occurred because of the flatness of the former cabinet that brought the aluminum case too close to the VFO tuned LC circuit. Aluminum has a huge tendency to expand under the influence of heat so the rig was very temperature sensitive. That undeniable fault lead to a complete reconstruction using the old RX and TX modules and building a new set of frequency generators.
Parts of the old cabinet were reused but because of the fact that the whole rig got increased vertical expansion, the cabinet was “stretched” with two lateral strips of Aluminum.
Also a full electronic transmit/receive switch with p-channel power MOSFETs has been designed to avoid usage of a DC switch relay and get a “smooth” switching.
Another objective of this radio was to get out the absolute best performing circuits of the recent projects and to build a real high-performance radio. Hence this transceiver is also some sort of an improvement of the “Old school SSB TRX” as well. The circuit empirically turned out to be very good for communication in the 14MHz band. Because of this frequent readers of this website might detect certain similarities. 😉
The Receiver section
The design objectives were:
Low noise (achieved by using Dual-Gate-MOSFETs with the receiver to a large extent)
High dynamic range (achieved by using a Dual-Gate-MOSFET as receive mixer)
High AGC range (achieved by taking RF preamp and IF amp into the AGC chain)
Good audio quality (achieved by using a TBA820M as integrated AF amplifier circuit and a 5 cm loudspeaker)
RF preamplifier and receive mixer
The radio frequency preamplifier has been designed primarily to improve the receiver’s noise figure. Delivering additional gain only is relevant in second order.
Preselection is performed with only one tuned circuit int G1 line. The center frequency of this circuit is 14.180MHz. In the output section of the stage an another identical LC circuit has been installed. This turned out to be sufficient because there is no immediate need of higher preselection. The subsequently placed mixer, that is also equipped with a Dual-Gate-MOSFET has very good high level processing qualities. No interfrequency feedthrough could be observed with various antennas. No IMD occured even when signals were very strong. Testing out in the field with partable antenna very far from man-made noise sources the receiver was very quiet and even very weak stations could be received and read with Q5.
To get most of gain swing from AGC the preamplifier is controlled by a DC voltage between 0 and 12V supplied by the AGC control stage to be described later. This voltage is halved by a 1:1 resistor voltage divider because maximum gain of the Dual-Gate-MOSFET occurs with about 6V DC applied to G2.
Clipping diodes that are sometimes used to prevent high voltage entering the 1st stage have not been installed because they are prone to produce unwanted IMD products if signal levels from the antenna are too high and undesired mixing takes place there.
To prevent self-oscillation in the preamplifier, the tuned circuit LC1 and LC2 are connected together in a special way. G1 is connected to the tuned section of LC1. This section has high impedance, thus it should be connected to a load which also has high impedance. The coupling section of the coil with low impedance is connected to the 50Ω antenna. There are not two tuned parts of the LC circuits together in one stage.
The output of the Dual-Gate-MOSFET (low impedance) is connected to the coupling winding, the high impedance tuned part is going the high impedance of G1 of the mixer. The impedance ratio between the two coils is 16:4 due to the winding ratio of 4:1 of the coil set.
The sensitivity and noise figure of the whole receiver is determined by these two stages. Measurements showed that the minimum discernable signal is about 0.1µV which is very good for a short wave receiver.
SSB-Filter, IF amplifier, Demodulator, AF amp and AGC
The following stages are some sort of best practice combination of circuits that have proven to perform very well in the recent projects.
SSB-Filter and relay
The SSB filter is switched with a special rf relay by Teledyne® ensuring excellent isolation of relay ports with very low capacities in the unswitched signal path. Here the usage of shielded cable is mandatory for connecting the relay/filter section to the transmitter (see later text!). A clamp diode has been installed to eliminate high voltage peaks due to self-induction when the relay is switched. This will prevent the MOSFETs in the switching unit from excessive voltage and possible destruction.
A proven and reliable circuit can be found here as well. One stage delivers IF gain of about 12dB which is sufficient because the mixer following as a demodulator (NE612) also propduces some dB of gain. Too much gain in this section only contributes to high noise in the speaker later and is not desirable.
The Dual-Gate-MOSFET in this stage is also integrated to the AGC chain. Together with the RF preamp installed in the front and also being part of AGC control end we will get some 20 to 25 dB of gain swing when AGC is fully driven. This turned out to be enough, only in some rare cases I found that the manual gain control (also included in this recevier) needs to be used in addition when AGC is not able to cope with excessive signal levels.
Compared to a MC1350 IC equipped IF amplifier I found that gain control is much smoother because the V->dB function is very much less precipitously with the Dual-Gate-MOSFET than it is with the MC1350.
NE612 is built-in here. The main advantage of this IC is that it requires only a few components and it has got an additional gain of about 12dB or more.
In VDD line you will find a 5.6V Zener to bring 12..14V supply voltage down to about 6V. There are also two capacitors. The 0.1uF is for bleeding off rf energy from or to the supply rail, the same is the purpose of the 10uF cap for audio frequencies or low frequency noise present on VDD line. This noise sometimes originates from the digital components in the radio and should be eliminated at all reasonable points in the circuit. Also it will help to prevent the high gain amplifier chain from self-oscillating in the audio frequency range.
Audio frequency amplifier section
Two ICs are used here. The first is an operational amplifier (uA741) with a 150kΩ resistor as part of negative feedback circuit. This value is comparatively low. If (in rare cases) higher gain should be needed it can be replaced by e. g. 330kΩ or even more.
The main audio amp is the TBA820M, an integrated audio amplifier in 8 pin DIL case. It is an interesting alternative for LM386 because tendency for self-ocillation is much lower within the TBA820M. But it requires some more components. TBA820M can be switched with the load (speaker) to VVD or GND. I use a headphone jack in the radio, that is grounded, hence I prefer the latter version.
A “good” loudspeaker with 5cm of diameter was found by ordering a larger series of different speakers from Chinese vendors via ebay. The differences in sound quality are breath-taking. So it is worthwhile spending some money and order a larger variety of speakers and install the very best one.
This is a circuit I have used many times and it has proven to work very reliable. If you wish to have different settings concerning attack and decay time then another cap can be added via a switch to GND in parallel to the 47uF cap. Another 100uF for example will give a few extra fractions of a second in attack/decay time.
A 20kΩ variable resistor is used for manual gain setting. The AGC voltage that is near to VDD (12V or more) is divided and so AGC and manual gain control can be combined. At least until the point where noch AGCing will take place because the resulting voltage is <3V.
The “AGC thres.” variable resistor shown in the schematic will determine the point where AGC becomes active. I usually set it that way that solely band noise does not affect the AGC. Stronger stations (coming with S5 or 6 with a commercial transceiver) should give first minor influence on the AGC voltage. That is the point where amp gain should start dropping gradually. Strong stations must set AGC voltage to nearly 0 V.
The Transmitter section
The transmitter generally consists of two parts:
The SSB generator and the TX mixer, and
the Power Amplifier.
The full schematic of the two parts together:
Starting from the left we see the microphone amp. A nostalgic but still available operational amplifier integrated circuit (741) is used here. The amp has high gain (about 30dB) to make a dynamic microphone connectable. There is no DC feeding for an electret microphone. If you should wish to use one then the negative feedback resistors should be lowered to about 47kΩ and the audio level should be carefully observed to avoid excessive driving. DC must also be applied for htis type of microphone!
Double sideband generator
The MC1496 (still available as NOS in 14 pin DIL case or fresh from the market in SMD package by ON Semiconductors) offers high carrier suppression of about 50 to 60 dB. Therefore a network of 2 x 10kΩ and a 50kΩ variable resistor has been installed. The crucial point: To make full usage of this network the carrier offset must be set properly. If you should notice that there is no point within the full swing of the 50kΩ variable resistor then the carrier frequency should be readjusted.
A balanced output transformer has been installed to improve carrier suppression and to enhance output voltage.
SSB filter coupling out
The usage of shielded cable is mandatory here to avoid transfer of rf stray energy into the DSB and SSB line!
This stage also is equipped with an NE612 doubly balanced mixer due to reasons of circuit simplicity.
14MHz Band pass filter
This filter also needs observation. I use the TOKO style coil formers familiar from other projects. The winding data can be found in the schematic. The coil formers must have the ferrite caps and metal shield cans on to avoid incoupling of rf energy from the subsequent power stages. The filter should be placed away from the higher power stages to avoid self-oscillation inside the transmitter section.
RF amplifier power stages
The amplifier presented here has been tested in 2 different radios so far and has proven to be very stable, very linear and very rugged against antenna mismatch. The power levels are about 10 db gain per stage. From the second stage on the output impedance is 50Ω. This makes it easier to measure power levels with a 50Ω standard dummy load.
The 2 watt driver stage uses a PI-filter instead of a broadband transformer. This is because I intended to save some space on the veroboard and for a monoband transmitter this is a practical solution. If you should find out that there is a mismatch that results in losing gain, then the capacitors can slightly be modified because the L-network has impedance transforming capabilities. By knowing input versus output impedance and calculating a “Q”-factor subsequently L and C can be computed to get a defined step-down impedance (Link for further information). This is a useful method and, in case of low pass filter like applied here, there is also a filter for harmonics.
Driver and PA power amp are biased for AB-mode, all other stages operate in A-mode to ensure best linearity. Strategies using emitter degeneration and negative feedback are inherent in preamp and predriver stage.
All transistors apart from preamp stage require usage of heat sinks.
Impedance matching is either not done (stage 1 to 2), by transformer (stage 2 to 3) or by L-network (stage 3 to 4). Whereas from stage 3 to 4 also there is a transformer applied to split the signal symmetrically to the two bases of the final transistors.
If there should be a tendency for self-oscillation within this stage the input transformer secondary winding can be center tapped and put to GND via a 0.1 capacitor.
Power out depends on DC power voltage and is about 20 watts when run on 13.5 V DC power supply an the amplifier terminated to a 50Ω load.
This is a spectroscopical analysis of the fully driven transmitter (f=14.200kHz, Pout = 20.1 watts, VDD=13.0V) and the remaining carrier.
Harmonics are filtered very effectively . This is achieved by using a push-pull final stage driven in AB mode. Some authors say this is useful to eliminate odd number harmonics. On the other hand there are two sections of low pass filtering (one between driver and PA, one following PA). The figure of the output spectrum between and 50 MHz below:
The Dual DDS Oscillator System
The DDS has got the following features:
VFO: AD9951 + amplifier,
LO: AD9834 + amplifier,
LCD: NOKIA 5110,
Tuning: Optical rotary encoder by Bourns,
User interface: 4 keys to control the digital settings,
Analog inputs: User keys (ADC0), VDD (ADC1), S-Value (ADC2), TX PWR (ADC3), PA Temp. (ADC4).
The control lines for DDS1 (AD9951) and DDS2 (Ad9834) are as follows:
The colors are the colors used for the cables in my radio.
The LCD is connected likewise:
The NOKIA5110 LCD has been designed for VDD=3.3V. Please use 10kΩ resistors in the control lines which are not in the schematic! 3.3V are derived more or less closely by switching 2 Si-Diodes in series which results in a voltage drop of about 1.4V. Hence the LCD gets 3.6V DC from the 5V supply chain which is no problem for the module. One big advantage of the Nokia LCD should not be forgotten: It is very quiet and does not produce any discernable digital noise. Thus it is my favourite meanwhile for receivers on the RF bands.
For both DDS modules coupling out the rf is done with symmetrical circuits using trifilar broadband transformers. 10 turns on a FT37-43 core are a good choice. This will enhance gain and reduce spurs.
DDS2 is clocked to 110MHz, but keep in mind, that AD9834 is specified for 75 MHz max. clock rate only. I found out that modules from the “grey market” sometimes fail and produce lousy signals when overclocked. You can see that on a scope when extra peaks appear or with the spectrum analyzer when spurious signal are frequent. I recommend buying with Mouser or anther trusty vendor for example or reduce clock rate in case of problems in signal quality.
Power consumption is not excessive because both DDS modules are for low power application, unlike the AD9850 or AD9835, that draw much higher current. Power rate is 300mA when in receive mode with LCD backlight on.
The C-code for the software has about 2600 lines source code and can be downloaded here.
A standard CB DC supply cable is used here. Unfortunatel the plugs equipped with a cable and fuse holder are widely availabe but the sockets have to be stripped from old CB trasnceivers.
On the air the transceiver performs great. Audio is clear and powerful what the QSO partners often tell me. The receiver is fun to listen to, sounding soft AND precise. Maybe I will do a YouTube video the next weeks to prove it! 😉
All my rigs are for portable, hiking, bicycle trips and travel to foreign countries. I use Aluminum as a basis for the hardware to keep the radio lightweight. With this radio a ground plane made of 0.8mm Aluminum sheet metal has been used that one has been enforced with a lateral additional ground plane carrying the DDS system (see pictures in this article, please). Thus the base frame is pretty rigid and not prone for bending.
Front an rear panel are made from 0,.8mm Al sheets (rear) and 0.5 mm Al sheet (front).
The various subassemblies (DDS, receiver, transmitter) are split into different modules and are seperatelay fixed with bolts and washers mounted to special spacer bolts for screws of 2mm diameter. This ensures better grounding instead of using larger veroboards. Connections are made from flexible stranded hook-up wire and shielded cable for rf and audio signals.
On the undersides of the single boards copper foil is used for lines with GND portential.
I am currently testing a remake of the “Old School SSB TRX” modified by adding a Dual-DDS-VFO to enhance frequency accuracy and stability. The rig is completed and a description will follow within the next weeks. But another finding is notable.
Currently I am using WEBSDR sites to monitor my own signal from time to time. On sdr.hu you can find a large number of WEBSDR sites from all over the world. Most of them are somehow insensitive because they seem to use suboptimal antennas. M0RZF appears to be different.
I first had an initial test monitoring the two-tone signal for some seconds on 14.200 MHz with the WEBSDR and the transceiver connected to a half wave dipole antenna.
After some seconds I switched the transmitter off, disconnected the antenna and connected a BNC cable terminated by a 50Ω dummy load. I watched the signal on the scope and the spectrum analyzer and did some measurements. Output power was about 20 watts PEP.
When adjusting the transmitter I, by accident, got the PC monitor from the corner of the eye with the SDR still on it. I was pretty surprised when I noticed this figure:
Well, the antenna cable is on the desk about half a meter distant from the dummy load and it seems that there is enough stray energy coupled into the 50Ω-antenna-cable to produce a discernable signal over 1000 kilometers away. But nonetheless, pretty surprising.
Conclusion: Always be careful when testing your transmitters with a dummy load what you are about to talk into the microphone! Use a two-tone test generator instead! 😉
When building a direct-digital-synthesis (DDS) frequency generator, the engineer has to take into account one inherit shortcoming of this state-of-the-art technology: Spurious signals that are an unwanted product of generating a radio frequency signal with a synthesizer. These signals occur in a wide range of frequencies and are a limiting factor for receiver performance particularly sensitivity. A method of examining and evaluating these unwanted signal products will be described and some guidelines for amplifier design will be presented.
Spurious signals in a DDS system originate from various signal sources in
the microcontrollers (MCU) driving the DDS. The responsible parts inside the MCU are clock oscillators, dividers, pulse-width-modulation (PWM) timers etc.
the DDS chip itself mainly from clock dividers and the digital-analog-converters (DACs) used.
Various factors contribute to the problem. First the topology of the synthesizer itself. These systems contain a DAC to form a sine wave signal out of a computer calculated synthesizing data model. DACs have a wide variety of bitwidth. A rule of the thumb is: The more bits the DAC has, the lower the number and the weaker the spurious signals will be. 14-bit DACs by ANALOG DEVICES e. g. perform sufficiently for low noise receivers.
Another topic is clock rate. The occurrence of unwanted signals out of the synthesis process is a reciprocal function of clock rate. So it is highly recommended to use the highest possible clock rate the respective chip is designed for. When experimenting with the AD9951 the following findings occured: From 200MHz primary clock the number of spurious signals significantly decreases. The usage of an internal clock multiplier (if available) is not recommended since it will deteriorate phase noise because another oscillator is added to the signal generating chain..
So far the theory that is common today. Another aspect should now be brought into discussion: The role of the stages that are the successors behind the mere synthesizer. First stage usually is an amplifier that is used to lift the signal level of the synthesizer (usually about 1Vpp.) to a level that it is needed for a certain type of mixer.
The DDS circuit
To eliminate any weakness in the basic generator underlying this research a high-level performance synthesizer has been constructed. This ensures a pure sine wave output which is essential because we want to examine the potentially negative outcomes of the various small signal amplifiers succeeding the synthesizer.
The DDS chip used in this circuit is the AD9951 by Analog Devices that incorporates a 14-bit digital-analog-converter (DAC). Clock rate for the chip here is 200MHz (400MHz max. according to data sheet), clock output is 1.8Vpp. which is the maximum signal level that is suitable for this 1.8V-technology based DDS.
The AD9951 DDS integrated circuit needs 2 supply voltages: 1.8V for the digital and analog circuits and 3.3V for DVDD_I/O, the output driver voltage.
Controls lines are 5V applicable which makes the DDS suitable for being controlled by a 5V microcontroller as well as a 3.3V system.
The signal outlet in this case is made from a symmetrical transformer (3 parallel windings 10 turns each on a FT43-37 core) using the IOUT1 and its corresponding paraphase outlet IOUT2. To use the balanced output is another effective possibility to reduce spurious signals as well as to enhance signal voltage by about 3 to 5 dB.
As first step the unamplified signal shall be inspected. In a wider spectroscopic range (f0=8.65MHz, fgen.=16MHz, f1=250.0MHz) the signal performs as shown below where f0 and f1 are the edge frequencies of the spectrum analyzer and fgen. is the output frequency of the DDS):
A first spurious signal can be detected with a signal level of more than 40dB below the generated signal. The signals at around 100MHz supposedly are strong FM stations in the VHF radio band whose energy from a nearby radio tower is coupled into the laboratory via the short wave antenna cable ending on top of the workbench. The peaks around 200 MHz are likely generated by the DDS clock oscillator.
Switching to a more narrow spectrum, we get this reading:
Remarkable that there is no even first harmonic that should be expected around the 32 MHz region. It is highly probable that this elimination of the 1st harmonic is caused by the symmetrical decoupling of the signal from the DDS. Hence we know that push-pull operated amplifiers reduce distortion and therefore tend to minimize even harmonics production.
Remarkable on the right falling edge of the main peak there another signal occurs hidden by the main curve which requires further examination.
Examining amplifiers for a DDS system
1.) Bipolar RF preamplifier circuit (adapted from DeMaw et. al., Solid state design for the Radio Amateur)
The first amplifier under test is a simple circuit containing a bipolar transistor. To reduce distortion emitter degeneration and negative feedback (from collector to base) have been installed. This is an amplifier that is often used in rf amplifiers as first stage of the power strip. Therefore it should contribute less to the overall distortion of the circuit.
Overall voltage gain with a 2SC829 RF transistor (fT=230MHz) is 13dB with 1 MHz, decreasing about 3 db per octave, power gain has not been evaluated.
With the settings of the spectrum analyzer unchanged it turns out that this amplifier obviously produces new signals that are prone to disturb the receiver of a radio where this amplifier is installed:
Signal level is about 2V pp.
One countermeasure is to carefully check the input level of the amplifier. Excessive input voltage will bring the amplifier into the clipping area thus generating IMD products and harmonics. We usually do not only observe the output signal with an oscilloscope but use the spectrum analyzer in parallel. This ensures optimized signal quality.
Proper biassing is essential for this type of amplifier, aside from the linearization described before. The operating point (also referred to as “Q-point) must be set in the middle of the linear part of the IBE->IC function.
Usually this is achieved by applying a positive voltage (for NPN transistor) so that a given “quiescent” current flow through the base-emitter line. In the most simple case a voltage divider with the base connected to the joint of the two resistors works satisfactory.
Any AC voltage applied now will alter the voltage sum of DC (quiescent) and AC around the Q-point.
Currently I am revising older projects that are in my radio shelf, some of them not finished yet, postponed to a later date, some without a cabinet, some with severe problems with performance and so on. All the stuff that needs a “second chance” ;-). This project is one of this collection. The transmitter did not work correctly (severe parasetic oscillations occurred when the section was driven to power levels >1 watt).
By careful testing and examining I found the reason: The grounding of the rf power amplifier stage was defective due to a connection that had not been soldered properly. After having cured that I found the output was 5 to 6 watts PEP output (very clean). Then, having the project on “GO!”, I finished the design. Thus I got a nice little “vintage style” SSB QRP trsanceiver as a travel or hiking companion:
Frequent readers on my blog know that one thing I really enjoy is building radios based on a minimalist concept. The fewer components you need for a working transceiver, the better it is. At least in my point of view. Here is another one of these “very lean design” transceivers.
The radio originally was designed as a study for my “Old School Transceiver“. After having not built a “real” analog VFO for a number of years I wanted to find out if I still can set up a construction that is really stable concerning frequency. And because it is not very challenging to just watch the result on a frequency counter, a full transceiver had to be built along with the VFO. The VFO was OK, (see later text!) the power transmitter, as mentioned before, was not. Until I had revised it.
The design is another remake of the „Kajman Transceiver“ by SQ7JHM. A design I absolutely love because of its simplicity. The radio basically has been designed for 80 meters (even when lot of websites quote it as a 20m rig) so it shows some weaknesses when adapted to 14MHz without any changes. Thus some improvements had to be made.
Improving performance of the SQ7JHM basic design
Some changes that were top of the agenda to meet my requirements:
The receiver needed a preamplifier for bands where atmospheric noise is not that strong. A dual-gate MOSFET equipped radio frequency preamplifier improves noise figure significantly and can be put into the AGC chain to give more dynamic range and a more pleasant listening experience.
An AGC (automatic gain control) is a good idea if you want to use the receiver in a more comfortable way without the need to lower the volume when strong stations appear. In addition the S-meter reading can be derived from the output of the AGC DC amplifier stage.
A little bit more rf output power can be achieved by using a push-pull amplifier. Linearity also improves to a certain degree when using this design because AB mode combined with separated amplification of the half waves plus suppression of even-order harmonics.
To enhance receiver gain a single stage interfrequency amplifier has been added that is only in use when on receive. It is also connected to the AGC chain.
And, last, a microphone amplifier allows you to talk in a moderate way into the microphone which is good for me because I often have my QSOs when the rest of the family is asleep and not keen on listening to my strange “This is DK7IH/QRP, do you copy?” messages.
The schematic of my enhanced design:
Fascination originates from the fact that you only need a handful of components (OK your hand should not have the size of that of a new born baby!) to set up a working short wave SSB transceiver.
Some thoughts on frequency stability
Careful design is the key for stable operation. This means component selection as well as setting it up on the veroboard.
The basic problem for every conventional free running VFO is temperature and its influence on the size of components. Due to the theory of thermodynamics all materials change their mechanical dimensions with temperature. This is caused by the kinetic energy of the molecules forming the crystals of a solid body. Thermal energy leads to enhanced oscillation of the molecules and therefore the need of larger spaces of each in individual molecule in a crystal. Because we have capacitors in a tuned circuit this will affect the values of all caps (wanted and unwanted ones) to a certain degree.
Something that helps the builder is called “temperature coefficient”. This means that electronic components increase OR decrease their respective value when they get warmer. The first is called “positive temperature coefficient”, the opposite is called “negative temperature coefficient”. So, you might guess, the fine art of radio building involves the knowledge of the characteristic behavior of components when heated.
I quote my findings about temperature behavior listed in the article referred to on the beginning of this text:
Ceramic capacitors: —
Polystyrene capacitor: –
NP0 (C0G) capacitor: no measurable effect
Air coil on polystyrene coil former: +++
Coil wound on T50-6 yellow toroid: +
The more “+” or “-” signs, the more steep the function of T->dC or T->dL is. So you can see: The best choice are polystyrene capacitors combined with coil on a yellow toroid. This combination is likely to outbalance temperature effects. If extra capacity is needed, NP0 caps are recommended.
From the existing principles of building a free running radio frequency oscillator I prefer the Hartley circuit. It uses a tapped coil (tap about 1/5 from the “lower” end) and saves capacitors compared to the Colpitts design. The tap achieves in-phase feedback. The lower you put the tap to the end the lower the amount of fed back energy will be. This leads to more frequency stability because the circuit does not heat up by excessive internal radio frequency. But be sure that oscillation is always strong enough and does not stop. The Hartley circuit is more simple and caps always inherit the risk of thermal problems when poorly selected.
The tuning is done with a Vernier drive and a homemade variable capacitor. For this a foil variable cap of an old AM radio has been dismantled an reassembled with air as dielectric. Lots of experiments were necessary to get the “frequency swing” correct and the basic capacitance to the right area.
Other measures that support frequency stability are :
Low DC power into the oscillator stage (avoids heating the device up by DC current),
Stabilizing voltage for the VFO stage by 2 consecutive steps,
Using a FET instead of bipolar transistor (no PN boundary layers in a FET),
Very loose coupling between oscillator and buffer stage reduce fed back of impedance changes by the output,
Low impedance output with emitter follower,
Avoid metal sheets (spec. Aluminum) close to the tuning elements! Aluminum sheet metal changes its size largely with even low temperature differences.
This oscillator is stable. It needs 5 to 10 minutes to settle which is in the normal range of what can be expected. I then can have it tuned to one frequency and there is a maximum change in frequency < 50Hz for hours. And, to compare with synthesizer technology: NO birdies at all. Really not. I love it! 😉
The mixers and filter section
NE602 and its derivatives have been used in legions of amateur transceivers. Basically designed for cell phones and small cordless phones radio amateurs quickly have found out that this mixer IC can be the universal mixer in lots of possible amateur radio designs. The main weakness is its low IMD3. But for a 14MHz rig the risk of appearance of strong out-of-band signals is not that likely. Besides, the selectivity of the receiver’s input section supports this. Strong in-band signals did not appear so far due to low band conditions. We’ll have to see how the receiver performs here.
On the other hand NE602 gives a good sensitivity which makes it ideal for radios on the higher bands where signal levels are not so high.
The NE602 has a balanced input AND a balanced output. This allows the designer to get two different signal sources to the input then subsequently mixed with the oscillator signal. As well the two outputs can be used to send the mixed signal to different paths.
This is what is the basic idea behind the design described here.
The mixer that is used together with the microphone to produce the DSB signal by mixing the audio signal with the local oscillator (LO) also serves as the product detector on receive by mixing the interfrequency with the LO. Correct signal path is set with the two relays depending on the fact you are either on transmit or receive mode.
The same principle is for the other mixer. It is transmit mixer or receive mixer, depending on the position of the relays.
The relays connect the SSB filter either to the input or the output of a distinct mixer. A graphical presentation should make it clear:
RX amp and interfrequency amplifier
These 2 stages are more or less the same. They provide 2 to 12 dB of gain depending on the AGC voltage applied to gate 2 of the dual gate MOSFET. In this version of the radio a potentiometer of 20kΩ is used to have the possibility to lower the DC voltage manually, by doing this an MGC (manual gain control) is achieved in a simple way.
A bipolar transistor and the inevitable LM386 amplify the filtered audio signal from the product detector to a volume that can be discerned even in a louder environment. The audio low pass filter prior to the AF preamp should be selected due to the users individual preferences concerning tone pitch of the audio signal.
RF power amplifier
This is more or less my standard power amplifier for small QRP rigs. I put stress on linear amplification, so I use emitter degeneration and negative feedback in collector circuit to get best IMD3 results. Even if the circuit could deliver one or two more watts I let the output power level at about 5 watts pep.
Here ist the result of a dual tone modulation:
Voltage division is 10 volts per cm, so this is 45Vpp which equals to about 5 watts max. peak output. Quite OK for QRP. And here is the spectrum of a 2-tone-modulated signal:
The whole transceiver is built on a 12×8 cm Veroboard (4.7″ x 3.1″). There is only one layer. The cabinet is 4 cm high (1.55″), 14 cm long (5.5″) and 9 cm wide (3.5″).
Left the vernier drive with the homemade capacitor attached. Left of the 9MHz filter you can see the LO, more far left the S-meter (from an old CB radio) hiding the audio amps. The 2 mixer ICs and the relays are sited around the SSB-filter. On the right side the power amp partly hidden by the DC switching board.
Well, that’s the story how a nearly failed project was saved from the scrapyard and came to life by carefully searching the faulty element in the circuit.