I have used a simple dipole for 14MHz temporarily. This antenna was not much as quiet as the Deltaloop but much simpler to install and fix, compared to the loop that has been my standard antenna for 14MHz for years. The noise in the receiver when using the dipole was beyond words.
Thus I changed the dipole for a quad loop again the last weekend. When connecting the transceiver I found that in spite of good SWR (1:1.2) the transmitter was self-oscillating. With the spectrum analyzer I could recognize that there were oscillations in the medium wave range as well as around 14 MHz. This made me examining the transmitter more closely.
After estimated a dozen of checks and failed attempts I changed to improve the performance I decided to alter the input transformer of the final stage. This item, which by origin had a winding ratio of 4 to 2+2, then was changed by a transformer having 3 to 3+3. This even deteriorated the situation.
Next I concluded that the second winding might have too high impedance. So I went to a 4 to 1+1 transformer (pig nose core BN43-202).
This finally cured the problem to a 100% and gave a very stable transmitter. Output power increased to 28 watts when fully modulated. This might be the direct outcome of better impedance matching.
Here is the current transmitter circuit with all the improvements made so far:
The first power amplifier for this transceiver project initially was capable to produce 20 watts of SSB pep rf power on the 20 meter band. The transistors in use are the 2SC1969 bipolar types by eleflow.com. A single device is rated to max. output power of 16 watts according to data sheet. In push-pull mode this nearly doubles because each of the semiconductors only has to amplify half of the duty cycle. Hence I started trying to get a little bit more power out of the PA assembly.
First I modified T3. The former data was: 1+1 turn primary center tapped and 4 turns secondary on a homemade pig nose core of 2×3 stacked FT50-43 toroids.
The new transformer has got 1 secondary turn more. Wire is 0.8mm enam.wire both for primary and secondary.
First I tried out 0.5mm enam. wire for the secondary which resulted in about 23 watts output pep. Next I used 0.8 mm diameter enam. wire which reduces the consequences of skin effect significantly . Penetration depth of a 14MHz ac current is only some 50 µm, thus every increase in surface area reduces resistance.
This set of measurements greatly improves two figures:
Output power increases from 20 to 25 watts pep,
carrier suppression improves by 5 dB.
The enhanced signal on the RIGOL DS1054 scope:
About 100 volts pp. (i. e. 35.5 volts rms) at 50Ω equals to about 25 watts pep. Nice improvement for a minor change that was done within 20 minutes!
A general and good practice in engineering is a steady process of improvement. This article describes the construction of a high performance transmitter/receiver for SSB (voice) communication covering the 14MHz (20 meters) high frequency amateur radio band.
Various modules that have proven high performance, liability and ruggedness in recent constructions will be combined to form a radio with outstanding receiver performance, an ultra linear transmitter with output range 15 to 20 watts and a top audio sound quality both on transmit and receive.
Key features are:
Dual DDS frequency generation with AD9834 (Local oscillator) and AD9951 (VFO),
Microcontroller (MCU): ATMega644P by ATMEL,
Single conversion superhet receiver with 9MHz interfrequency (IF) and preamplifier, mixer and IF amplifier equipped with Dual-Gate-MOSFETs,
Audio-derived automatic gain control (AGC),
Transmitter with MC1496 as double sideband (DSB) modulator and NE602 as transmit mixer,
power transimitter with 4 stages, final stage in push-pull mode.
Another version of this radio has been built before. But this one was equipped with a variable frequency oscillator (VFO) because of nostalgia reasons. Unfortunately a VFO lacks certain features (frequency stability above all) which can be overcome by using digital frequency synthesis without losing performance. Usage of a high performance DDS systems is a prerequisite to achievement and a possible solution.
Most building blocks of that respective radio have been redesigned except the VFO section that turned out as not being able to deliver the projected frequency stability to a 100% degree. Frequency instability occurred because of the flatness of the former cabinet that brought the aluminum case too close to the VFO tuned LC circuit. Aluminum has a huge tendency to expand under the influence of heat so the rig was very temperature sensitive. That undeniable fault lead to a complete reconstruction using the old RX and TX modules and building a new set of frequency generators.
Parts of the old cabinet were reused but because of the fact that the whole rig got increased vertical expansion, the cabinet was “stretched” with two lateral strips of Aluminum.
Also a full electronic transmit/receive switch with p-channel power MOSFETs has been designed to avoid usage of a DC switch relay and get a “smooth” switching.
Another objective of this radio was to get out the absolute best performing circuits of the recent projects and to build a real high-performance radio. Hence this transceiver is also some sort of an improvement of the “Old school SSB TRX” as well. The circuit empirically turned out to be very good for communication in the 14MHz band. Because of this frequent readers of this website might detect certain similarities. 😉
The Receiver section
The design objectives were:
Low noise (achieved by using Dual-Gate-MOSFETs with the receiver to a large extent)
High dynamic range (achieved by using a Dual-Gate-MOSFET as receive mixer)
High AGC range (achieved by taking RF preamp and IF amp into the AGC chain)
Good audio quality (achieved by using a TBA820M as integrated AF amplifier circuit and a 5 cm loudspeaker)
RF preamplifier and receive mixer
The radio frequency preamplifier has been designed primarily to improve the receiver’s noise figure. Delivering additional gain only is relevant in second order.
Preselection is performed with only one tuned circuit int G1 line. The center frequency of this circuit is 14.180MHz. In the output section of the stage an another identical LC circuit has been installed. This turned out to be sufficient because there is no immediate need of higher preselection. The subsequently placed mixer, that is also equipped with a Dual-Gate-MOSFET has very good high level processing qualities. No interfrequency feedthrough could be observed with various antennas. No IMD occured even when signals were very strong. Testing out in the field with partable antenna very far from man-made noise sources the receiver was very quiet and even very weak stations could be received and read with Q5.
To get most of gain swing from AGC the preamplifier is controlled by a DC voltage between 0 and 12V supplied by the AGC control stage to be described later. This voltage is halved by a 1:1 resistor voltage divider because maximum gain of the Dual-Gate-MOSFET occurs with about 6V DC applied to G2.
Clipping diodes that are sometimes used to prevent high voltage entering the 1st stage have not been installed because they are prone to produce unwanted IMD products if signal levels from the antenna are too high and undesired mixing takes place there.
To prevent self-oscillation in the preamplifier, the tuned circuit LC1 and LC2 are connected together in a special way. G1 is connected to the tuned section of LC1. This section has high impedance, thus it should be connected to a load which also has high impedance. The coupling section of the coil with low impedance is connected to the 50Ω antenna. There are not two tuned parts of the LC circuits together in one stage.
The output of the Dual-Gate-MOSFET (low impedance) is connected to the coupling winding, the high impedance tuned part is going the high impedance of G1 of the mixer. The impedance ratio between the two coils is 16:4 due to the winding ratio of 4:1 of the coil set.
The sensitivity and noise figure of the whole receiver is determined by these two stages. Measurements showed that the minimum discernable signal is about 0.1µV which is very good for a short wave receiver.
SSB-Filter, IF amplifier, Demodulator, AF amp and AGC
The following stages are some sort of best practice combination of circuits that have proven to perform very well in the recent projects.
SSB-Filter and relay
The SSB filter is switched with a special rf relay by Teledyne® ensuring excellent isolation of relay ports with very low capacities in the unswitched signal path. Here the usage of shielded cable is mandatory for connecting the relay/filter section to the transmitter (see later text!). A clamp diode has been installed to eliminate high voltage peaks due to self-induction when the relay is switched. This will prevent the MOSFETs in the switching unit from excessive voltage and possible destruction.
A proven and reliable circuit can be found here as well. One stage delivers IF gain of about 12dB which is sufficient because the mixer following as a demodulator (NE612) also propduces some dB of gain. Too much gain in this section only contributes to high noise in the speaker later and is not desirable.
The Dual-Gate-MOSFET in this stage is also integrated to the AGC chain. Together with the RF preamp installed in the front and also being part of AGC control end we will get some 20 to 25 dB of gain swing when AGC is fully driven. This turned out to be enough, only in some rare cases I found that the manual gain control (also included in this recevier) needs to be used in addition when AGC is not able to cope with excessive signal levels.
Compared to a MC1350 IC equipped IF amplifier I found that gain control is much smoother because the V->dB function is very much less precipitously with the Dual-Gate-MOSFET than it is with the MC1350.
NE612 is built-in here. The main advantage of this IC is that it requires only a few components and it has got an additional gain of about 12dB or more.
In VDD line you will find a 5.6V Zener to bring 12..14V supply voltage down to about 6V. There are also two capacitors. The 0.1uF is for bleeding off rf energy from or to the supply rail, the same is the purpose of the 10uF cap for audio frequencies or low frequency noise present on VDD line. This noise sometimes originates from the digital components in the radio and should be eliminated at all reasonable points in the circuit. Also it will help to prevent the high gain amplifier chain from self-oscillating in the audio frequency range.
Audio frequency amplifier section
Two ICs are used here. The first is an operational amplifier (uA741) with a 150kΩ resistor as part of negative feedback circuit. This value is comparatively low. If (in rare cases) higher gain should be needed it can be replaced by e. g. 330kΩ or even more.
The main audio amp is the TBA820M, an integrated audio amplifier in 8 pin DIL case. It is an interesting alternative for LM386 because tendency for self-ocillation is much lower within the TBA820M. But it requires some more components. TBA820M can be switched with the load (speaker) to VVD or GND. I use a headphone jack in the radio, that is grounded, hence I prefer the latter version.
A “good” loudspeaker with 5cm of diameter was found by ordering a larger series of different speakers from Chinese vendors via ebay. The differences in sound quality are breath-taking. So it is worthwhile spending some money and order a larger variety of speakers and install the very best one.
This is a circuit I have used many times and it has proven to work very reliable. If you wish to have different settings concerning attack and decay time then another cap can be added via a switch to GND in parallel to the 47uF cap. Another 100uF for example will give a few extra fractions of a second in attack/decay time.
A 20kΩ variable resistor is used for manual gain setting. The AGC voltage that is near to VDD (12V or more) is divided and so AGC and manual gain control can be combined. At least until the point where noch AGCing will take place because the resulting voltage is <3V.
The “AGC thres.” variable resistor shown in the schematic will determine the point where AGC becomes active. I usually set it that way that solely band noise does not affect the AGC. Stronger stations (coming with S5 or 6 with a commercial transceiver) should give first minor influence on the AGC voltage. That is the point where amp gain should start dropping gradually. Strong stations must set AGC voltage to nearly 0 V.
The Transmitter section
The transmitter generally consists of two parts:
The SSB generator and the TX mixer, and
the Power Amplifier.
The full schematic of the two parts together:
Starting from the left we see the microphone amp. A nostalgic but still available operational amplifier integrated circuit (741) is used here. The amp has high gain (about 30dB) to make a dynamic microphone connectable. There is no DC feeding for an electret microphone. If you should wish to use one then the negative feedback resistors should be lowered to about 47kΩ and the audio level should be carefully observed to avoid excessive driving. DC must also be applied for htis type of microphone!
Double sideband generator
The MC1496 (still available as NOS in 14 pin DIL case or fresh from the market in SMD package by ON Semiconductors) offers high carrier suppression of about 50 to 60 dB. Therefore a network of 2 x 10kΩ and a 50kΩ variable resistor has been installed. The crucial point: To make full usage of this network the carrier offset must be set properly. If you should notice that there is no point within the full swing of the 50kΩ variable resistor then the carrier frequency should be readjusted.
A balanced output transformer has been installed to improve carrier suppression and to enhance output voltage.
SSB filter coupling out
The usage of shielded cable is mandatory here to avoid transfer of rf stray energy into the DSB and SSB line!
This stage also is equipped with an NE612 doubly balanced mixer due to reasons of circuit simplicity.
14MHz Band pass filter
This filter also needs observation. I use the TOKO style coil formers familiar from other projects. The winding data can be found in the schematic. The coil formers must have the ferrite caps and metal shield cans on to avoid incoupling of rf energy from the subsequent power stages. The filter should be placed away from the higher power stages to avoid self-oscillation inside the transmitter section.
RF amplifier power stages
The amplifier presented here has been tested in 2 different radios so far and has proven to be very stable, very linear and very rugged against antenna mismatch. The power levels are about 10 db gain per stage. From the second stage on the output impedance is 50Ω. This makes it easier to measure power levels with a 50Ω standard dummy load.
The 2 watt driver stage uses a PI-filter instead of a broadband transformer. This is because I intended to save some space on the veroboard and for a monoband transmitter this is a practical solution. If you should find out that there is a mismatch that results in losing gain, then the capacitors can slightly be modified because the L-network has impedance transforming capabilities. By knowing input versus output impedance and calculating a “Q”-factor subsequently L and C can be computed to get a defined step-down impedance (Link for further information). This is a useful method and, in case of low pass filter like applied here, there is also a filter for harmonics.
Driver and PA power amp are biased for AB-mode, all other stages operate in A-mode to ensure best linearity. Strategies using emitter degeneration and negative feedback are inherent in preamp and predriver stage.
All transistors apart from preamp stage require usage of heat sinks.
Impedance matching is either not done (stage 1 to 2), by transformer (stage 2 to 3) or by L-network (stage 3 to 4). Whereas from stage 3 to 4 also there is a transformer applied to split the signal symmetrically to the two bases of the final transistors.
If there should be a tendency for self-oscillation within this stage the input transformer secondary winding can be center tapped and put to GND via a 0.1 capacitor.
Power out depends on DC power voltage and is about 20 watts when run on 13.5 V DC power supply an the amplifier terminated to a 50Ω load.
This is a spectroscopical analysis of the fully driven transmitter (f=14.200kHz, Pout = 20.1 watts, VDD=13.0V) and the remaining carrier.
Harmonics are filtered very effectively . This is achieved by using a push-pull final stage driven in AB mode. Some authors say this is useful to eliminate odd number harmonics. On the other hand there are two sections of low pass filtering (one between driver and PA, one following PA). The figure of the output spectrum between and 50 MHz below:
The Dual DDS Oscillator System
The DDS has got the following features:
VFO: AD9951 + amplifier,
LO: AD9834 + amplifier,
LCD: NOKIA 5110,
Tuning: Optical rotary encoder by Bourns,
User interface: 4 keys to control the digital settings,
Analog inputs: User keys (ADC0), VDD (ADC1), S-Value (ADC2), TX PWR (ADC3), PA Temp. (ADC4).
The control lines for DDS1 (AD9951) and DDS2 (Ad9834) are as follows:
The colors are the colors used for the cables in my radio.
The LCD is connected likewise:
The NOKIA5110 LCD has been designed for VDD=3.3V. Please use 10kΩ resistors in the control lines which are not in the schematic! 3.3V are derived more or less closely by switching 2 Si-Diodes in series which results in a voltage drop of about 1.4V. Hence the LCD gets 3.6V DC from the 5V supply chain which is no problem for the module. One big advantage of the Nokia LCD should not be forgotten: It is very quiet and does not produce any discernable digital noise. Thus it is my favourite meanwhile for receivers on the RF bands.
For both DDS modules coupling out the rf is done with symmetrical circuits using trifilar broadband transformers. 10 turns on a FT37-43 core are a good choice. This will enhance gain and reduce spurs.
DDS2 is clocked to 110MHz, but keep in mind, that AD9834 is specified for 75 MHz max. clock rate only. I found out that modules from the “grey market” sometimes fail and produce lousy signals when overclocked. You can see that on a scope when extra peaks appear or with the spectrum analyzer when spurious signal are frequent. I recommend buying with Mouser or anther trusty vendor for example or reduce clock rate in case of problems in signal quality.
Power consumption is not excessive because both DDS modules are for low power application, unlike the AD9850 or AD9835, that draw much higher current. Power rate is 300mA when in receive mode with LCD backlight on.
The C-code for the software has about 2600 lines source code and can be downloaded here.
A standard CB DC supply cable is used here. Unfortunatel the plugs equipped with a cable and fuse holder are widely availabe but the sockets have to be stripped from old CB trasnceivers.
On the air the transceiver performs great. Audio is clear and powerful what the QSO partners often tell me. The receiver is fun to listen to, sounding soft AND precise. Maybe I will do a YouTube video the next weeks to prove it! 😉
All my rigs are for portable, hiking, bicycle trips and travel to foreign countries. I use Aluminum as a basis for the hardware to keep the radio lightweight. With this radio a ground plane made of 0.8mm Aluminum sheet metal has been used that one has been enforced with a lateral additional ground plane carrying the DDS system (see pictures in this article, please). Thus the base frame is pretty rigid and not prone for bending.
Front an rear panel are made from 0,.8mm Al sheets (rear) and 0.5 mm Al sheet (front).
The various subassemblies (DDS, receiver, transmitter) are split into different modules and are seperatelay fixed with bolts and washers mounted to special spacer bolts for screws of 2mm diameter. This ensures better grounding instead of using larger veroboards. Connections are made from flexible stranded hook-up wire and shielded cable for rf and audio signals.
On the undersides of the single boards copper foil is used for lines with GND portential.
Currently I am revising older projects that are in my radio shelf, some of them not finished yet, postponed to a later date, some without a cabinet, some with severe problems with performance and so on. All the stuff that needs a “second chance” ;-). This project is one of this collection. The transmitter did not work correctly (severe parasetic oscillations occurred when the section was driven to power levels >1 watt).
By careful testing and examining I found the reason: The grounding of the rf power amplifier stage was defective due to a connection that had not been soldered properly. After having cured that I found the output was 5 to 6 watts PEP output (very clean). Then, having the project on “GO!”, I finished the design. Thus I got a nice little “vintage style” SSB QRP trsanceiver as a travel or hiking companion:
Frequent readers on my blog know that one thing I really enjoy is building radios based on a minimalist concept. The fewer components you need for a working transceiver, the better it is. At least in my point of view. Here is another one of these “very lean design” transceivers.
The radio originally was designed as a study for my “Old School Transceiver“. After having not built a “real” analog VFO for a number of years I wanted to find out if I still can set up a construction that is really stable concerning frequency. And because it is not very challenging to just watch the result on a frequency counter, a full transceiver had to be built along with the VFO. The VFO was OK, (see later text!) the power transmitter, as mentioned before, was not. Until I had revised it.
The design is another remake of the „Kajman Transceiver“ by SQ7JHM. A design I absolutely love because of its simplicity. The radio basically has been designed for 80 meters (even when lot of websites quote it as a 20m rig) so it shows some weaknesses when adapted to 14MHz without any changes. Thus some improvements had to be made.
Improving performance of the SQ7JHM basic design
Some changes that were top of the agenda to meet my requirements:
The receiver needed a preamplifier for bands where atmospheric noise is not that strong. A dual-gate MOSFET equipped radio frequency preamplifier improves noise figure significantly and can be put into the AGC chain to give more dynamic range and a more pleasant listening experience.
An AGC (automatic gain control) is a good idea if you want to use the receiver in a more comfortable way without the need to lower the volume when strong stations appear. In addition the S-meter reading can be derived from the output of the AGC DC amplifier stage.
A little bit more rf output power can be achieved by using a push-pull amplifier. Linearity also improves to a certain degree when using this design because AB mode combined with separated amplification of the half waves plus suppression of even-order harmonics.
To enhance receiver gain a single stage interfrequency amplifier has been added that is only in use when on receive. It is also connected to the AGC chain.
And, last, a microphone amplifier allows you to talk in a moderate way into the microphone which is good for me because I often have my QSOs when the rest of the family is asleep and not keen on listening to my strange “This is DK7IH/QRP, do you copy?” messages.
The schematic of my enhanced design:
Fascination originates from the fact that you only need a handful of components (OK your hand should not have the size of that of a new born baby!) to set up a working short wave SSB transceiver.
Some thoughts on frequency stability
Careful design is the key for stable operation. This means component selection as well as setting it up on the veroboard.
The basic problem for every conventional free running VFO is temperature and its influence on the size of components. Due to the theory of thermodynamics all materials change their mechanical dimensions with temperature. This is caused by the kinetic energy of the molecules forming the crystals of a solid body. Thermal energy leads to enhanced oscillation of the molecules and therefore the need of larger spaces of each in individual molecule in a crystal. Because we have capacitors in a tuned circuit this will affect the values of all caps (wanted and unwanted ones) to a certain degree.
Something that helps the builder is called “temperature coefficient”. This means that electronic components increase OR decrease their respective value when they get warmer. The first is called “positive temperature coefficient”, the opposite is called “negative temperature coefficient”. So, you might guess, the fine art of radio building involves the knowledge of the characteristic behavior of components when heated.
I quote my findings about temperature behavior listed in the article referred to on the beginning of this text:
Ceramic capacitors: —
Polystyrene capacitor: –
NP0 (C0G) capacitor: no measurable effect
Air coil on polystyrene coil former: +++
Coil wound on T50-6 yellow toroid: +
The more “+” or “-” signs, the more steep the function of T->dC or T->dL is. So you can see: The best choice are polystyrene capacitors combined with coil on a yellow toroid. This combination is likely to outbalance temperature effects. If extra capacity is needed, NP0 caps are recommended.
From the existing principles of building a free running radio frequency oscillator I prefer the Hartley circuit. It uses a tapped coil (tap about 1/5 from the “lower” end) and saves capacitors compared to the Colpitts design. The tap achieves in-phase feedback. The lower you put the tap to the end the lower the amount of fed back energy will be. This leads to more frequency stability because the circuit does not heat up by excessive internal radio frequency. But be sure that oscillation is always strong enough and does not stop. The Hartley circuit is more simple and caps always inherit the risk of thermal problems when poorly selected.
The tuning is done with a Vernier drive and a homemade variable capacitor. For this a foil variable cap of an old AM radio has been dismantled an reassembled with air as dielectric. Lots of experiments were necessary to get the “frequency swing” correct and the basic capacitance to the right area.
Other measures that support frequency stability are :
Low DC power into the oscillator stage (avoids heating the device up by DC current),
Stabilizing voltage for the VFO stage by 2 consecutive steps,
Using a FET instead of bipolar transistor (no PN boundary layers in a FET),
Very loose coupling between oscillator and buffer stage reduce fed back of impedance changes by the output,
Low impedance output with emitter follower,
Avoid metal sheets (spec. Aluminum) close to the tuning elements! Aluminum sheet metal changes its size largely with even low temperature differences.
This oscillator is stable. It needs 5 to 10 minutes to settle which is in the normal range of what can be expected. I then can have it tuned to one frequency and there is a maximum change in frequency < 50Hz for hours. And, to compare with synthesizer technology: NO birdies at all. Really not. I love it! 😉
The mixers and filter section
NE602 and its derivatives have been used in legions of amateur transceivers. Basically designed for cell phones and small cordless phones radio amateurs quickly have found out that this mixer IC can be the universal mixer in lots of possible amateur radio designs. The main weakness is its low IMD3. But for a 14MHz rig the risk of appearance of strong out-of-band signals is not that likely. Besides, the selectivity of the receiver’s input section supports this. Strong in-band signals did not appear so far due to low band conditions. We’ll have to see how the receiver performs here.
On the other hand NE602 gives a good sensitivity which makes it ideal for radios on the higher bands where signal levels are not so high.
The NE602 has a balanced input AND a balanced output. This allows the designer to get two different signal sources to the input then subsequently mixed with the oscillator signal. As well the two outputs can be used to send the mixed signal to different paths.
This is what is the basic idea behind the design described here.
The mixer that is used together with the microphone to produce the DSB signal by mixing the audio signal with the local oscillator (LO) also serves as the product detector on receive by mixing the interfrequency with the LO. Correct signal path is set with the two relays depending on the fact you are either on transmit or receive mode.
The same principle is for the other mixer. It is transmit mixer or receive mixer, depending on the position of the relays.
The relays connect the SSB filter either to the input or the output of a distinct mixer. A graphical presentation should make it clear:
RX amp and interfrequency amplifier
These 2 stages are more or less the same. They provide 2 to 12 dB of gain depending on the AGC voltage applied to gate 2 of the dual gate MOSFET. In this version of the radio a potentiometer of 20kΩ is used to have the possibility to lower the DC voltage manually, by doing this an MGC (manual gain control) is achieved in a simple way.
A bipolar transistor and the inevitable LM386 amplify the filtered audio signal from the product detector to a volume that can be discerned even in a louder environment. The audio low pass filter prior to the AF preamp should be selected due to the users individual preferences concerning tone pitch of the audio signal.
RF power amplifier
This is more or less my standard power amplifier for small QRP rigs. I put stress on linear amplification, so I use emitter degeneration and negative feedback in collector circuit to get best IMD3 results. Even if the circuit could deliver one or two more watts I let the output power level at about 5 watts pep.
Here ist the result of a dual tone modulation:
Voltage division is 10 volts per cm, so this is 45Vpp which equals to about 5 watts max. peak output. Quite OK for QRP. And here is the spectrum of a 2-tone-modulated signal:
The whole transceiver is built on a 12×8 cm Veroboard (4.7″ x 3.1″). There is only one layer. The cabinet is 4 cm high (1.55″), 14 cm long (5.5″) and 9 cm wide (3.5″).
Left the vernier drive with the homemade capacitor attached. Left of the 9MHz filter you can see the LO, more far left the S-meter (from an old CB radio) hiding the audio amps. The 2 mixer ICs and the relays are sited around the SSB-filter. On the right side the power amp partly hidden by the DC switching board.
Well, that’s the story how a nearly failed project was saved from the scrapyard and came to life by carefully searching the faulty element in the circuit.
This article describes the “Cigarette Pack” SSB QRP transceiver” for 14MHz that I first had mentioned some months before. Recently, when taking it from the shelf, the transceiver dropped to the floor and was severely damaged. This lead to serious defects in the front panel area, the main frame, the cabinet and so on. The interior parts were, luckily, not affected by the crash. So, I had to revise the whole radio, make a new front panel and cabinet, ply the frame straightly (as far as possible) and so on. This is the full description of the rig now to complete the files here. The good news: The radio is fine again and fully operational! And the even better news: I still have not started smoking!
During reconstruction the transceiver has been extended for about 5 mm so that overall length now is 100mm (3.9 inch). This was done because I intended to build in a loudspeaker. The other dimensions remain unchanged: Width is 52mm (2 inch.), height is 30mm (1.2inch). OK it is slightly longer now than a standard pack of cancer sticks, but who cares? Total cabinet volume is 150cm³.
The transceiver is based on the “Micro 23” rig, that I have described here. Some simplifications of that already simplified radio have been made. Here is the full schematic of this even smaller transceiver:
Very simple rigs like this one always use parts of the circuit for receive and transmit purpose. Here these parts are the 2 mixers (NE602), the SSB-filter and the interfrequency amplifier.
Signal flow schematic
The NE602 has a balanced output. With mixer 1 only one of them is used. If higher gain is desired, a broadband (or even better a tuned LC circuit) transformer could be used to connect pin 4 and 5 (the mixer outputs) in push-pull mode. I did not do that to save the transformer.
The signal flow can be derived from the design:
Receive mode signal flow
From the antenna relay (not drawn) the rf energy runs through a 2 pole LC filter for 14 MHz. The coils are wound small TOKO coil formers, all respective data is given in the schematic. Coupling is loose via a 3.3pF cap.
NExt stage is an rf preamp for 14MHz with a broadband output. The acitve element here is a dual-gate MOSFET.
After having left this stage the 14MHz signal travels through another 470pF capacitor. This one has high resistance for audio frequency and low for rf frequencies due to the equation: XC =1/(2*PI*f*C). The signal is then fed, together with the audio signal from the microphone (when on transmit), into mixer 1 input on pin 1. The 1k resistor prevents the rf energy from flowing into the microphone circuit. The two signals are separated from each other by simply exploiting reactance and resistance in a rather clever way 😉.
When receiving the Si5351A clock chip is programmed in a way that the VFO signal (23 MHz) is present on output CLK0. It is fed into mixer 1 via a small cap to prevent overloading of the mixer. The Si5351A breakout board delivers about 3 Vpp. clock signal, so this must be reduced to about 200mVpp. A 5.6pF capacitor is OK here.
The resulting signal is sent to the SSB filter (a 9MXF24D) that is terminated with 1kOhm and 20pF in parallel. The wanted SSB signal is present at the output of the filter.
Next stage is the interfrequency amplifier, equipped with a dual-gate-MOSFET semiconductor. This one is connected to the AGC chain, on receive a variable voltage is applied to gate 2 (range 0 to 6 V), on transmit the AGC is fully powered to ensure maximum gain.
Next is mixer 2 which is the product detector when receiving. The signal (9MHz +/- sideband shift) is applied to pin 6. Due to the fact that this mixer also serves as transmit mixer, the two signals are taken from the two mixer outputs on pin 4 (serving as audio output) and pin 5 (serving as rf output for transmitting).
Two audio amplifiers (preamplifier and power stage) give a sufficient signal level for an 8 ohm loudspeaker or a headphone.
For the loudspeaker I tried out the tiny ones for smartphones with good success. Only the volume was a little bit low. Then I found another speaker in an old toy of my daughter that turned out to be very much OK for this transceiver. Its diameter is about 3 cm (1.2 inch) and just fits in the housing.
Transmit signal flow
The microphone in this radio is an electret one. The advantage is that these microphones have an internal preamplifier equipped with a field-effect-transistor. The output voltage is fairly high, about 1Vpp. when normally speaking into it. Therefore an audio preamp is obsolete. The microphone signal is directly fed into pin 1 of the first mixer. On transmit the Si5351 signal generator is switched that the 9MHz (+/- sideband shift) signal is fed into pin 6. The SSB filter eliminates the unwanted sideband, the interfrequency amplifier lifts the SSB signal to an appropriate level. The TX mixer is fed with the 23MHz signal resulting in a 14 and 37 MHz signal. The TX band pass filter cleans the signal from the unwanted 37MHz component resulting from the mixer process.
RF power amplifier
The power amplifier is a 3 stage circuit. Stage 1 (preamplifier) brings the signal to about 10 mW. This is coupled into the driver stage via a cap of 0.1uF without any further impedance matching.
The subsequent driver stage shifts the signal level to about 200mW. Linear amplification is ensured her (as well as in the previous stage) by negative feedback in the collector circuit and emitter degeneration with a non-bypassed resistor to GND. An output transformer (winding rate 4:1, impedance rate thus 16:1) lowers the impedance of some 100 ohms to a few 10 ohms present on the input of the final amplifier stage.
The final amplifier brings up a signal level of 3 to 4 Watts PEP. This stage is in AB mode, the appropriate bias is achieved by the 1k resistor going to +12V TX and the current to GND via the silicon diode. This diode must be thermally connectod to the final transistor to stabilize the bias.When the transistor heats up, the silicon diode increases the current through it thus decreasing bias to the transistor.
The 68 ohm resistors serves 2 purposes: First it prevents the input signal from being shorted by the bypass caps in the bias circuit and it stabilizes the rf behavior of the stage by limiting the gain because certain amounts of the input power are led to GND. This prevents self-oscillation.
DC ad the collector is fed through a radio frequency choke to hinder rf from flowing into the DC line. Radio frequency is directly fed into the low-pass-filter. The output impedance of this stage is roughly 50 Ohms, so the filter can be a 50 ohm circuit with a cutoff frequency slightly above 14MHz.
The VFO section
The Si5351A clock chip used here has three frequency outputs that can be set individually. Only CLK0 and CLK1 are used in this radio. The Si5351A chip is programmed by software in the following manner:
Receive: CLK0 is the VFO, CLK1 is the BFO.
Transmit: CLK0 is the BFO, CLK1 is the VFO.
The microcontroller reads the tx/rx status and switches the frequencies respectively.
The radio is a full SMD design on a 0.1″ pitch double sided Veroboard:
The control panel on the left with tuning knob and volume set. The 64×32 pixel OLED between these controls. Following the microcontroller behind the fron panel (here covered). The controller is an ATmega168 on an Arduino Pro mini board.
The isolated board left of the SSB is the AGC section. The receiver and transmitter shared parts follow, the TX band pass filter is in the foreground. The power transmitter is on the right behind the shield. The shield is necessary to avoid unwanted oscillations when rf is coming back from the power transmitter to the band pass filter prior to the tx section.
On the right there is the SMA socket for connecting the antenna plus a 3 pin header for connecting a headphone. When there is no headphone in use a jumper connects the internal speaker to the speaker line. VDD is applied via a standard DC connector.
The underside of the board has only some SMD components and the wiring on it:
“On the air”
My longest distance achieved with this transceiver (after rebuilding it) has been R2DLS near Moscow who gave me a “59”-report. The antenna in use is, as always, a Deltaloop.
Hi again! This project directly “beams” you back to the “Good ol’ 80s” when there was no stuff like “DDS, “OLED” or even “SDR” or other modern technology we today use to build our radios.
I designed this transceiver using the “old school” techniques because in a German QRP forum on the internet some hams originated a “Back to the roots”-movement which I thought was a great idea. So I too went back in time 3 decades and constructed a radio like I did it in the eighties at the beginning of my “homebrew career”. That meant: No digital stuff, just a simple VFO but (and that is new) higher rf output power because condx are fairly low on the hf bands currently.
I later presented this radio at an annual German convention of homebrewing hams called the “Black Forest Meeting” named by the place where it is held the beginning of October each year.
To give you an impression, that’s how the radio looks from the outside. Pretty “old school”, isn’t it?
The main design objectives were very simple:
Compact in size (even without using SMD components),
Analog VFO with vernier drive (1:10 gear) and variable capacitor,
No digital stuff (=> no digital noise!),
RF Output in the range from 15 to 20 watts pep in SSB,
Single conversion superhet (9MHz interfrequency)
No “save as many components as possible”-design.
First the block diagram giving you the basic structure of this radio::
“Old school” SSB transceiver for 14MHz by DK7IH (2018) – basic outline
I decided to use an analog VFO in this project due to three reasons:
It’s really old style,
it is much less prone to produce any unwanted “birdies”, and
phase noise performance usually is better than most of the digital ways to generate a signal.
For the VFO I chose the Hartley design characterized by a tapped coil. This type uses less critical components than a comparable Colpitts circuit thus reducing number of parts (2 caps in this case that are avoided) which might lead to unwanted frequency changes (drift).
How to build a VFO that is really stable
Lots of pages have been written about this topic. This another one. First, be aware of the fact that it is not possible to build a VFO that has the same frequency stability like a modern digital system. This is because these systems are all crystal controlled. But it is possible to achieve a drift of some dozen Hertz within an hour or so which is absolutely sufficient for having even a longer QSO.
The main problem is based on physics, or thermodynamics to say more exactly. All material expands when heated and contracts when ambient temperature decreases. OK, some exceptions exist, water below 4°C is the commonly known example of them.
Avoiding thermal runaway
Heat is the problem in such a circuit. It comes from the interior of the components when current flows through them and from the outside, for example when the transceiver is exposed to sunlight or placed near another source of thermal energy. Also heating of the final rf amplifier stages may contribute to heating the cabinet inside. The electronic parts forming the central strucure of the tuned circuit exert the main influence connected to thermal runaway of the frequency that is generated.
The general approach is: When we can’t avoid physical effects we must choose components that change their values in such a way to compensate the changing of the values of the other parts. That means we have to look carefully on temperature coefficients of the varoius components we intend to build into our VFO.
Choosing the “right” components for your VFO
Choosing advantageous components is crucial for frequency stability. So I did some brief research to find out more about temperature coefficients of coils of various types and available capacitors. Here are some of the outcomes.
Explanation of syntax: If a relation is negative, a minus sign (“-“) is given. In this case the value (C or L) decreases when temperaure increases. A plus sign (“+”) indicates a positive coefficient. When the relation of value change by temperature change is weak (that means no intense changing of the value when heated), there is only one “-“-sign. The more “-“-signs you have, the higher this respective ratio is. The same applies for “+”-signs to indicate a positive relation.
Ceramic capacitors: —
Polystyrene capacitor: –
NP0 (C0G) capacitor: no measurable effect
Air coil on polystyrene coil former: +++
Coil wound on T50-6 yellow toroid: +
Based on this short survey, the best combination would be NP0- and Polystyrene caps together with an inductor wound on an T50-6 (yellow) core. Hopefully their temperature behavoiur will compensate more or less and lead to best stability. Hint: On the photos appearing later in this text you will see an air coil wound on a TOKO style coil former that has been used because it does not need so much space.
The VFO circuit
I finally chose the Hartley circuit for my VFO. There it is:
VFO Circuit explanation
Starting from the left you can see the tapped coil (here 60 turns tapped at 10 turns from the bottom end) on a 5.5 mm TOKO style coil former without any core. In parallel there are various capacitors (polystyrene and NP0 mixed) to build up the total capcity. It is common use to spread the total capacity needed to various single capacitors because it has turned out that the effects of temperature change are less significant if you use more (and therefore smaller) single capacitors.
A 100k resistor is used to pull the gate to ground and therefore provides a correct bias at the FET’s gate. The 1N914 diode is a so called “clamp” diode that has been installed to stabilize (and therefore reduce) the rf voltage in order to avoid excessive rf voltage coming to the FET’S gate which would lead to distortion. This diode has a negative side effect, but that an be accepeted for a VFO in the rf bands: It slightly increases phase noise because it works as a regulator. With some the designs you can see this diode in reverse position, don’t worry, the regulating effect takes place either.
To ensure the oscillator to produce radio waves, in-phase feedback between gate and source is generated via the tap you can see with the coil. A tap of about 1/6 of the whole number of windings provides enough feedback voltage to let the oscillator start by its inherent thermal noise and generate clear sine waves afterwards. Putting the tap too close to the “hot end” will cause distortion because the amount of energy coupled back to the gate will be too high. Also instabilites are probable because of excessive drive power to the gate of the FET.
On top of the tuned circuit there is a varactor diode that is used to be controlled by a positive voltage to form a RIT (receiver independend tuning) control circuit. It is very loosely coupled to the tuned circuit to minimze temperature effects and because only 1 or 2 kHz “swing” is needed. The generation of the RIT voltage will be described later in this text.
The main tuning capacitor
An air capacitor is mandatory here! You can either use a ready made one from the surplus market. But to keep it as small as possible I built my own by dismanteling one of old variable capacitors formerly used for homemade AM radios. Use a small drill to remove the rivets, dismantle the capacitor completely and rebuild it again as an air capacitor (get rid of the plastic dielectric interlayers!) by using M2-screws and nuts. Youl will have evenings of endless fun with this game! 😉
Buffering and amplifying the Signal
The second stage with another FET is very loosely coupled to the source of the first FET. This is made to minimize effects of load changes to the frequency. This stage is a so called “source follower” giving a very low impedance signal to the final stage that is responsible for the amplification of the signal to a level of 2 to 3 volts pp which you will need for the rx mixer that has been designed as a dual-gate-MOSFET mixer (see receiver chapter later!).
DC voltage in the VFO
Voltage stabilization is crucial for best performance of this critical part of the radio. Supply voltage changes always lead to frequency changes. So a two-level buffering is common use here. The first (and most critical) stage is buffered twice (10V voltage regulator integrated circuit 78L10 and subsequently by a 6.2V zener diode) whereas the buffer and the amplifier stages are supplied with 10V regulated DC voltage only.
Ambient thermal isolation
To avoid the VFO being affected by interior thermal convection (flow of warm air inside the cabinet) it is recommended to shield the VFO from the rest of the transceiver. I do not recommend using metal sheets as walls here because these form other unwanted capacities that will lead to thermal effects on the generated frequency. Metal also is a good conductor for thermal energy, so you might run counter to your goals. My thermal insulation therefore is made of simple cardboard.
The Local Oscillator (LO)
This oscillator is much more uncritical than the VFO because it is crystal controlled. The purpose of the LO is to supply a carrier signal for the SSB modulator. Due to the fact that there are two sidebands we theoretically can use this LO must be switched to either one of two possible frequencies. In case of an interfrequency of 9MHz (9000kHz) these are: 9001.5 kHz for the first sideband and 8998.5 kHz for the second sideband. Please note that I did not write “USB” or “LSB” because the frequencies forming each sideband might be changed because of the frequency plan of the transceiver where by mixing with the VFO frequency the sideband might be inverted depending on if you add or substract the VFO frequency from the 9MHz-SSB signal.
There are several possibilities to produce these two frequencies:
Using two different oscillators each equipped with a single crystal,
switching two crystals with one osillator,
using a variabale capacitor or a coil to “pull” one crystal to the desiered frequency.
This method means high effort but surely is the most exact one because there are no influences of the unneeded choice to the other crystal currently on duty.
Is the worst idea because the unswitched crystal is highly prone to influence the freqeuncy of the switched one because they are linked to parasetic capacities within the wiring, the switch and so on. Forget this one espacially when using the internal oscillator of a NE602/SA612!
This to my point of view is the best compromise between circuit simplicity and function. You can see this way of sideband switching in my transceiver.
This is my local oscillator:
It is a simple Colpitts circuit where in-phase feedback and feedback voltage control are achieved by a series of two identical capacitors. A simple switch, a capacitor (90pF max.) and a coil (4 to 8uH max.) that are either connected to the base of the transistor via the 9MHz crystal determine the sidband freqeuncy of the oscillator. Signal is taken out via the collector.
The receiver will be presented step-by-step starting with the front end stage:
The RF Preamp
This stage is connected to the antenna relay. It provides an amount of basic amplification for the antenna signals. But that is not the main purpose. Noise figure improves significantly if you use a stage with low inherent noise. Thus a dual-.gate MOSFET is installed here. This semiconductor is also used to control stage gain because gate 2 of the MOSFET is connected to the AGC chain of the transceiver. About 12 dB gain swing are possible here. Stage gain is about 15dB.
Note the position of the primary and secondary windings of the input and the output coil. To avoid self-oscillation the output (drain) of the MOSFET is connected to the untuned part of the LC circuit. Coils must be shielded and should be equipped with ferrite heads (in the photo the piece in left bottom corner).
The Receiver’s Mixer
In this stage also a dual-gate MOSFET is used. This type of mixers provides good capabilities to cope with high signal levels without producing unwanted signals (high IMD3), gives some dB of gain and is low-noise also.
One “disadvantage”, if you want to say so, is that it needs a little bit of higher VFO drive (about 2 to 3 volts pp). Gate 2 bias is generated via the voltage drop on the source line. The tuned circuit in drain line is adjusted to the desired interfrequency. See the schematic for the exact winding data and parallel capacitor.
The SSB Filter section
Transmitter and receiver share the same SSB filter in this transceiver. So some sort of switching is recommended even if circuits exist that go without one. I used a high quality relay made by Teledyne that I bought in a 10-piece bundle for low price (1€ each!) via a well-known internet marketplace. Caution: Some SMD-relay I tested prior to building this rig were disastrous concerning signal isolation between terminals. To avoid any disappointment or frustration I recommend testing a relay before you finally install it.
All connections to the rest of the circuit must be made with shielded cable. I found an interesting alternative: I sometimes design my own very thin shielded “cable” with brass tubing (1 mm inside diameter) where I put insulated cable inside. The brass tube is connected to GND on the Veroboard. You can not bend these tubes but longer lines can be interrupted for a short piece so that the “bend” can be made by putting two parts of tubing in 90° degree angle for example.
The IF Amplifier
This one might look familiar to you. It is a simple “remake” of the front-end stage. The one remarkable thing is the secondary of the output transformer. This coil has 4 windings (prim. 16 turns). The secondary is center tapped (2 + 2 turns). This is because the product detector (SSB demodulator) has a symmetric input. Very important in this stage is the 100uF capacitor in VDD line. This cap prevents the stage from AF resonating and self-oscillations on the VDD line and makes the receiver much more “quiet”.
The SSB demodulator
This stage is probably the most “old-school” part of the whole transceiver. It uses an old CA3028A differential amplifier as mixer circuit:
You won’t be able to buy large amounts of this IC anymore. And if you get one, the prices are close to or beyond a rip-off. But there is an alternative. You can build your own “IC”. Watch this page where all the information you need is provided!
Signal input goes to the paired transistors forming the amplifier stages. LO is fed into the line transistor that is used to set the current of the differential amplifier thus providing a switching and therefore superposition of the two signals.
The output circuit is made of an audio transformer formerly used in the final audio amp of old AM radios (coil resistance is about 300Ω each side). The 2.2nF capacitors eliminate remainders of the rf signals and “ground” the terminals of the AF transformer.
This final receiver part consists of two stages: An audio preamplifier with a bipolar transistor and a final amp with a TBA820M integrated circuit.
The two caps 0.22 and 0.1uF determine how the higher frequency components in the audio signal are cut off. The higher the total value the more the higher frequency components of the audio signal will be limited due to the equation XC=1/(2*PI*f*C).
Tr1, which is a universal purpose NPN transistor, provides high gain. Thus a 10k resistor is installed to form a voltage divider with the audio gain potentiometer.
In the final stage I use a TBA820M ic (8-pin DIL version). This one is more linear than the well-known LM386 that you usually can find in this place and it is not so prone to self-oscillation. The cap aside the 100uF in the top left corner of the schematic is not marked, its value is 0.1uF.
Loudspeaker impedance is 8Ω.
Automatic gain control makes listening to signals much more comfortable. AGC voltage is audio derived, like in my other transceivers. The circuit also is nearly the same:
Due to the very high gain of the product detector this stage is directly connected to this circuit and not to the AF preamp. A potentiometer is used to set the threshold of the AGC onset.
Next stage is a simple audio preamplifier followed by a “Greinacher Circuit” serving as voltage doubler and providing DC voltage proportional to the audio signal level. A dc amplifier with another NPN transistor lets its collector voltage drop as soon as it is fed with significant dc input voltage. Thus this voltage decreases and so it can be used to control gate 2 of the MOSFETs in the various receiver stages that are equipped with tetrodes.
The S-Meter is connected to the emmitter of the final transistor. If conductivity in the transistor rises, the emmitter becomes more positive and the S-Meter needle is deflected proportionally. The 220Ω potentiometer in the emmitter line must be set in accordance to the respective S-Meter you are using. One shortcoming should be mentioned: If you have a not so sensitive meter then the value of the pot can be set to nearly 100Ω or above. This will prevent the collector from dropping to nearly 0V in case there is a strong signal and hence reduce the maximum dB you can get from the AGC chain.
The transmitter section is designed for an output level of about 20 watts and uses 4 stages all equipped with bipolar transistors. The last stage is a push-pull stage, the 3 low-power stages are single ended. I prefer push-pull for the last stage (if possible) because this circuit inherently does not create even harmonics thus simplifying output filtering.
The first parts of the transmitter to be shown here are the microphone amplifier, the SSB-generator and the TX mixer:
The mic amp is simple but provides enough gain and good linearity for using an old-style dynamic microphone. It works in common emmitter mode and has gain of about 15 to 20 dB.
The audio signal amplified by the microphone amplifier is fed into PIN1 (Input 1) of an NE602/SA612 mixer IC which is the simpliest way to generate a DSB signal with a Gilbert cell. LO input is fed to PIN7 and should be in the range of 200 to 300mVpp. Thus a 12pF cap has been installed to limit LO voltage going to input at PIN7.
Carrier suppression is around 45dB when LO offset frequency is correctly set for each of the two sidebands and LO voltage is not much higher than the 300mVpp mentioned before.
The DSB signal produced by this mixer goes on to the SSB filter relay and filter that has been described before. The use of shielded cable is mandatory, too.
The TX Mixer
You won’t be able to recognize many differences if you compare this TX mixer to the DSB generator. In fact, there are none.
The 14MHz Band Pass Filter
Next is the band pass filter that consists of 2 coupled tuned LC circuits for 14MHz. They are also wound on TOKO style coil formers. Data can be found in the schematic underneath.
It is important to also install the ferrite heads that are provided with most of the coil formers and to use the shield “metal cans” that are also standard for these coils. This is to prevent stray coupling of rf energy into the first stage of the power amplifier strip and therefore preventing self-oscillation of the transmitter strip.
For proper adjustment set the transmit frequency to about half of the frequency swing ((i. e. to about 14.200 kHz) and tune for max. output.
If you modulate with a two-tone signal to the mic amp you should see about 500mVpp by the output of the BPF when the chain is fully driven.
We start with the low power end of the power transmitter section. A bipolar rf type transistor is the center part of this stage.
This one is a standard circuit and has been “trimmed” for maximum linearity in order to reduce distortion to a minimum (which is also true for the following stages). You can see the well understood 2 master ways of achieving max. linearity in an amplifier stage:
Negative feedback between collector and base (i)
Emmitter degeneration (II)
i) The first measure goes along with the 2.7kΩ resistor between collector and base of the transistor. This resistor provides positive dc bias voltage to the base and leads 90° out-of-phase ac voltage to the transistor’s input. This reduces gain and therefore distortion. But due to the fact that the whole transmitter strip has plenty of gain, this loss in gain is not a serious problem.
ii) The 10Ω resistor in the emmitter line is not bypassed by a capacitor. This stabilizes the circuit. When the current through transistor increases the emmitter voltage will rise (according to Ohm’s law) and the voltage between collector and emmitter drops. This reduces voltage difference between base and emmitter and hence also reduces gain.
The coupling to the next stage is done by a capacitor of 0.1uF. This causes some impedance mismatch. But that is as well not a big problem because the gain reduction here helps to prevent the whole transmitter from unwanted oscillations by diminishing overall gain.
This stage is somehow a copy of the stage before but allows more current to flow through the stage. It is also operated in class “A” mode and uses the same methods to maximize linearity like the preamp stage.
You can use a 2N3866 transistor here which is available. But any other rf power transistor for driver stages (2SC1973 etc.) will also do the job well. A heatsink is recommended even if stage current ist not that high. T1 should be a toroid, a “pig-nose” core in this place to my experience is not the best choice. The 10uH RFCs are ready made ones but you can also wind 20 turns of 0.4mm enameled wire to a FT37-43 toroid core.
RF output of this stage could be measured as 100mW into a 50Ω load.
The Main Driver
This stage has an old 2SC2078 CB transistor and is operated in class “AB” mode. An alternative could be a 2N3553 that is available on ebay for example. A heatsink is neccessary for whatever type you use.
Correctly set bias for “AB” operation is ensured by the 1kΩ resistor from VDD to the bias circuit. The 1kΩ resistor limits the current whereas the diode works as a stabilizing element (thermistor). It must be connected directly to the case of the transistor ensuring good thermal contact. If the temperature of the devices rises the resistance of the diode will decrease. Hence current through the diode increases thus reducing the part of the current that can pass through the base-emmiter line of the 2SC2078. Quiescent current is stabilized and thermal runaway is prevented.
The rf output is uncommonly terminated with a low-pass-filter. This is because I first intended to build the transceiver for an output level of about 4 watts. But then I had the idea that the space still available on the veroboard could be used by another amplifier definitely leaving the QRP power level. So I left the circuit how it first was and just added the final amplifier stage.
Output of this driver stage now ist set to 1 watt into a 50Ω resistor.
The Final RF Amplifier
Now let’s go for the power machine in this transceiver:
2 rf power transistors 2SC1969 by Eleflow provide up to 20 watts of rf power. Bias for such a high power stage can not be set by a simple resistor. Here a line transistor (BD137) serves as current control. Diodes D1 and D2 (1N4002 or equ.) follow the same purpose like the single one in the stage described before. They must be mounted with excellent thermal contact to each of the 2 power devices which ensures secured protection against thermal runaway. The transistors also must be connected to a large heatsink. I use Aluminium metal strips (2mm thickness) to connect them to the back wall of the cabinet.
RF is fed into the power transistors via a network of 8.2Ω resistors and two 22uH rf chokes that seperate the rf line from the dc bias line letting only dc pass. This method makes construction of the input transformer easier. Winding ratio is 4 turns primary, 2 turns secondary. This is because the input impedance of the stage ist fairly low (aorund some ohms).
The output transformer is a homemade “pig-nose” of 6 toroids FT50-43, where 3 toroids are stacked (using 2-component glue) and 2 of these stacks are glued in parallel (see picture at the end of this text for details!). Winding ratio is 1 + 1 (primary center tapped) to 4 on secondary.
Quiescent current of this stage should be set to about 100mA.
A low-pass-filter terminates this stage and is connected to the antenna relay.
In addition you find a section to measure rf power. This is again the so called “Greinacher-Circuit” which doubles the voltage and serves as a charge pump. The dc output of this circuit directly leads to the S-Meter indicating output power of the transmitter.
First the spectrum of the signal with the transmitter fully driven to 20 watts output power with a two-tone-signal:
IMD3 is about 28dB below signal peak which I think is acceptable.
Amplitude diagram is as follows:
Max. radio frequency voltage is 90.4Vpp which calculates to about 20 watts of rf power (P=(Vpp/(2*SQR(2)))²/50Ω).
Power switch board and RIT voltage
A 12V relay with two pairs of contact sets is the heart of this unit. DC power is lead to TX, RX and permanent supply via the respective power lines.
RIT voltage generation is a little bit more complicated. When the RIT switch is in “OFF” position, RIT voltage always is taken from the fixed voltage divider that is formed of the two 4.7k resistors either when on receive or transmit mode.
If RIT is “ON” then there are two possibilities: When on receive mode, RIT voltage is gained from the 10k lin. potentiometer in the front panel. When on transmit mode RIT again is taken from the fixed voltage divider.
There is also a false polarity protection diode. This can be any silicon type with max. current >= 5 A.
The construction is sandwich style made of 2 layers:
OK, that’s the story. Thanks for joining me on the trip to the past! 😉
The challenge started some weeks ago, when John, ZL2TCA, commented to this blog
you next challange is to build a rig into a cigerette packet size case.
My problem: I don’t smoke, have never smoked and probably never will. 😉 But I have a new transceiver for 20 meters, that might come close to the dimensions of a pack of “cancer sticks”.
The transceiver is nearly the same circuit as applied with the “Micro 20-III” but uses a single ended final amplifier instead of a push-pull circuit. I hope to find time the next days to publish an article on this rig featuring full description of the radio. Currently I’m in the IOTA contest and working stations from all over Europe.