I have used a simple dipole for 14MHz temporarily. This antenna was not much as quiet as the Deltaloop but much simpler to install and fix, compared to the loop that has been my standard antenna for 14MHz for years. The noise in the receiver when using the dipole was beyond words.
Thus I changed the dipole for a quad loop again the last weekend. When connecting the transceiver I found that in spite of good SWR (1:1.2) the transmitter was self-oscillating. With the spectrum analyzer I could recognize that there were oscillations in the medium wave range as well as around 14 MHz. This made me examining the transmitter more closely.
After estimated a dozen of checks and failed attempts I changed to improve the performance I decided to alter the input transformer of the final stage. This item, which by origin had a winding ratio of 4 to 2+2, then was changed by a transformer having 3 to 3+3. This even deteriorated the situation.
Next I concluded that the second winding might have too high impedance. So I went to a 4 to 1+1 transformer (pig nose core BN43-202).
This finally cured the problem to a 100% and gave a very stable transmitter. Output power increased to 28 watts when fully modulated. This might be the direct outcome of better impedance matching.
Here is the current transmitter circuit with all the improvements made so far:
A general and good practice in engineering is a steady process of improvement. This article describes the construction of a high performance transmitter/receiver for SSB (voice) communication covering the 14MHz (20 meters) high frequency amateur radio band.
Various modules that have proven high performance, liability and ruggedness in recent constructions will be combined to form a radio with outstanding receiver performance, an ultra linear transmitter with output range 15 to 20 watts and a top audio sound quality both on transmit and receive.
Key features are:
Dual DDS frequency generation with AD9834 (Local oscillator) and AD9951 (VFO),
Microcontroller (MCU): ATMega644P by ATMEL,
Single conversion superhet receiver with 9MHz interfrequency (IF) and preamplifier, mixer and IF amplifier equipped with Dual-Gate-MOSFETs,
Audio-derived automatic gain control (AGC),
Transmitter with MC1496 as double sideband (DSB) modulator and NE602 as transmit mixer,
power transimitter with 4 stages, final stage in push-pull mode.
Another version of this radio has been built before. But this one was equipped with a variable frequency oscillator (VFO) because of nostalgia reasons. Unfortunately a VFO lacks certain features (frequency stability above all) which can be overcome by using digital frequency synthesis without losing performance. Usage of a high performance DDS systems is a prerequisite to achievement and a possible solution.
Most building blocks of that respective radio have been redesigned except the VFO section that turned out as not being able to deliver the projected frequency stability to a 100% degree. Frequency instability occurred because of the flatness of the former cabinet that brought the aluminum case too close to the VFO tuned LC circuit. Aluminum has a huge tendency to expand under the influence of heat so the rig was very temperature sensitive. That undeniable fault lead to a complete reconstruction using the old RX and TX modules and building a new set of frequency generators.
Parts of the old cabinet were reused but because of the fact that the whole rig got increased vertical expansion, the cabinet was “stretched” with two lateral strips of Aluminum.
Also a full electronic transmit/receive switch with p-channel power MOSFETs has been designed to avoid usage of a DC switch relay and get a “smooth” switching.
Another objective of this radio was to get out the absolute best performing circuits of the recent projects and to build a real high-performance radio. Hence this transceiver is also some sort of an improvement of the “Old school SSB TRX” as well. The circuit empirically turned out to be very good for communication in the 14MHz band. Because of this frequent readers of this website might detect certain similarities. 😉
The Receiver section
The design objectives were:
Low noise (achieved by using Dual-Gate-MOSFETs with the receiver to a large extent)
High dynamic range (achieved by using a Dual-Gate-MOSFET as receive mixer)
High AGC range (achieved by taking RF preamp and IF amp into the AGC chain)
Good audio quality (achieved by using a TBA820M as integrated AF amplifier circuit and a 5 cm loudspeaker)
RF preamplifier and receive mixer
The radio frequency preamplifier has been designed primarily to improve the receiver’s noise figure. Delivering additional gain only is relevant in second order.
Preselection is performed with only one tuned circuit int G1 line. The center frequency of this circuit is 14.180MHz. In the output section of the stage an another identical LC circuit has been installed. This turned out to be sufficient because there is no immediate need of higher preselection. The subsequently placed mixer, that is also equipped with a Dual-Gate-MOSFET has very good high level processing qualities. No interfrequency feedthrough could be observed with various antennas. No IMD occured even when signals were very strong. Testing out in the field with partable antenna very far from man-made noise sources the receiver was very quiet and even very weak stations could be received and read with Q5.
To get most of gain swing from AGC the preamplifier is controlled by a DC voltage between 0 and 12V supplied by the AGC control stage to be described later. This voltage is halved by a 1:1 resistor voltage divider because maximum gain of the Dual-Gate-MOSFET occurs with about 6V DC applied to G2.
Clipping diodes that are sometimes used to prevent high voltage entering the 1st stage have not been installed because they are prone to produce unwanted IMD products if signal levels from the antenna are too high and undesired mixing takes place there.
To prevent self-oscillation in the preamplifier, the tuned circuit LC1 and LC2 are connected together in a special way. G1 is connected to the tuned section of LC1. This section has high impedance, thus it should be connected to a load which also has high impedance. The coupling section of the coil with low impedance is connected to the 50Ω antenna. There are not two tuned parts of the LC circuits together in one stage.
The output of the Dual-Gate-MOSFET (low impedance) is connected to the coupling winding, the high impedance tuned part is going the high impedance of G1 of the mixer. The impedance ratio between the two coils is 16:4 due to the winding ratio of 4:1 of the coil set.
The sensitivity and noise figure of the whole receiver is determined by these two stages. Measurements showed that the minimum discernable signal is about 0.1µV which is very good for a short wave receiver.
SSB-Filter, IF amplifier, Demodulator, AF amp and AGC
The following stages are some sort of best practice combination of circuits that have proven to perform very well in the recent projects.
SSB-Filter and relay
The SSB filter is switched with a special rf relay by Teledyne® ensuring excellent isolation of relay ports with very low capacities in the unswitched signal path. Here the usage of shielded cable is mandatory for connecting the relay/filter section to the transmitter (see later text!). A clamp diode has been installed to eliminate high voltage peaks due to self-induction when the relay is switched. This will prevent the MOSFETs in the switching unit from excessive voltage and possible destruction.
A proven and reliable circuit can be found here as well. One stage delivers IF gain of about 12dB which is sufficient because the mixer following as a demodulator (NE612) also propduces some dB of gain. Too much gain in this section only contributes to high noise in the speaker later and is not desirable.
The Dual-Gate-MOSFET in this stage is also integrated to the AGC chain. Together with the RF preamp installed in the front and also being part of AGC control end we will get some 20 to 25 dB of gain swing when AGC is fully driven. This turned out to be enough, only in some rare cases I found that the manual gain control (also included in this recevier) needs to be used in addition when AGC is not able to cope with excessive signal levels.
Compared to a MC1350 IC equipped IF amplifier I found that gain control is much smoother because the V->dB function is very much less precipitously with the Dual-Gate-MOSFET than it is with the MC1350.
NE612 is built-in here. The main advantage of this IC is that it requires only a few components and it has got an additional gain of about 12dB or more.
In VDD line you will find a 5.6V Zener to bring 12..14V supply voltage down to about 6V. There are also two capacitors. The 0.1uF is for bleeding off rf energy from or to the supply rail, the same is the purpose of the 10uF cap for audio frequencies or low frequency noise present on VDD line. This noise sometimes originates from the digital components in the radio and should be eliminated at all reasonable points in the circuit. Also it will help to prevent the high gain amplifier chain from self-oscillating in the audio frequency range.
Audio frequency amplifier section
Two ICs are used here. The first is an operational amplifier (uA741) with a 150kΩ resistor as part of negative feedback circuit. This value is comparatively low. If (in rare cases) higher gain should be needed it can be replaced by e. g. 330kΩ or even more.
The main audio amp is the TBA820M, an integrated audio amplifier in 8 pin DIL case. It is an interesting alternative for LM386 because tendency for self-ocillation is much lower within the TBA820M. But it requires some more components. TBA820M can be switched with the load (speaker) to VVD or GND. I use a headphone jack in the radio, that is grounded, hence I prefer the latter version.
A “good” loudspeaker with 5cm of diameter was found by ordering a larger series of different speakers from Chinese vendors via ebay. The differences in sound quality are breath-taking. So it is worthwhile spending some money and order a larger variety of speakers and install the very best one.
This is a circuit I have used many times and it has proven to work very reliable. If you wish to have different settings concerning attack and decay time then another cap can be added via a switch to GND in parallel to the 47uF cap. Another 100uF for example will give a few extra fractions of a second in attack/decay time.
A 20kΩ variable resistor is used for manual gain setting. The AGC voltage that is near to VDD (12V or more) is divided and so AGC and manual gain control can be combined. At least until the point where noch AGCing will take place because the resulting voltage is <3V.
The “AGC thres.” variable resistor shown in the schematic will determine the point where AGC becomes active. I usually set it that way that solely band noise does not affect the AGC. Stronger stations (coming with S5 or 6 with a commercial transceiver) should give first minor influence on the AGC voltage. That is the point where amp gain should start dropping gradually. Strong stations must set AGC voltage to nearly 0 V.
The Transmitter section
The transmitter generally consists of two parts:
The SSB generator and the TX mixer, and
the Power Amplifier.
The full schematic of the two parts together:
Starting from the left we see the microphone amp. A nostalgic but still available operational amplifier integrated circuit (741) is used here. The amp has high gain (about 30dB) to make a dynamic microphone connectable. There is no DC feeding for an electret microphone. If you should wish to use one then the negative feedback resistors should be lowered to about 47kΩ and the audio level should be carefully observed to avoid excessive driving. DC must also be applied for htis type of microphone!
Double sideband generator
The MC1496 (still available as NOS in 14 pin DIL case or fresh from the market in SMD package by ON Semiconductors) offers high carrier suppression of about 50 to 60 dB. Therefore a network of 2 x 10kΩ and a 50kΩ variable resistor has been installed. The crucial point: To make full usage of this network the carrier offset must be set properly. If you should notice that there is no point within the full swing of the 50kΩ variable resistor then the carrier frequency should be readjusted.
A balanced output transformer has been installed to improve carrier suppression and to enhance output voltage.
SSB filter coupling out
The usage of shielded cable is mandatory here to avoid transfer of rf stray energy into the DSB and SSB line!
This stage also is equipped with an NE612 doubly balanced mixer due to reasons of circuit simplicity.
14MHz Band pass filter
This filter also needs observation. I use the TOKO style coil formers familiar from other projects. The winding data can be found in the schematic. The coil formers must have the ferrite caps and metal shield cans on to avoid incoupling of rf energy from the subsequent power stages. The filter should be placed away from the higher power stages to avoid self-oscillation inside the transmitter section.
RF amplifier power stages
The amplifier presented here has been tested in 2 different radios so far and has proven to be very stable, very linear and very rugged against antenna mismatch. The power levels are about 10 db gain per stage. From the second stage on the output impedance is 50Ω. This makes it easier to measure power levels with a 50Ω standard dummy load.
The 2 watt driver stage uses a PI-filter instead of a broadband transformer. This is because I intended to save some space on the veroboard and for a monoband transmitter this is a practical solution. If you should find out that there is a mismatch that results in losing gain, then the capacitors can slightly be modified because the L-network has impedance transforming capabilities. By knowing input versus output impedance and calculating a “Q”-factor subsequently L and C can be computed to get a defined step-down impedance (Link for further information). This is a useful method and, in case of low pass filter like applied here, there is also a filter for harmonics.
Driver and PA power amp are biased for AB-mode, all other stages operate in A-mode to ensure best linearity. Strategies using emitter degeneration and negative feedback are inherent in preamp and predriver stage.
All transistors apart from preamp stage require usage of heat sinks.
Impedance matching is either not done (stage 1 to 2), by transformer (stage 2 to 3) or by L-network (stage 3 to 4). Whereas from stage 3 to 4 also there is a transformer applied to split the signal symmetrically to the two bases of the final transistors.
If there should be a tendency for self-oscillation within this stage the input transformer secondary winding can be center tapped and put to GND via a 0.1 capacitor.
Power out depends on DC power voltage and is about 20 watts when run on 13.5 V DC power supply an the amplifier terminated to a 50Ω load.
This is a spectroscopical analysis of the fully driven transmitter (f=14.200kHz, Pout = 20.1 watts, VDD=13.0V) and the remaining carrier.
Harmonics are filtered very effectively . This is achieved by using a push-pull final stage driven in AB mode. Some authors say this is useful to eliminate odd number harmonics. On the other hand there are two sections of low pass filtering (one between driver and PA, one following PA). The figure of the output spectrum between and 50 MHz below:
The Dual DDS Oscillator System
The DDS has got the following features:
VFO: AD9951 + amplifier,
LO: AD9834 + amplifier,
LCD: NOKIA 5110,
Tuning: Optical rotary encoder by Bourns,
User interface: 4 keys to control the digital settings,
Analog inputs: User keys (ADC0), VDD (ADC1), S-Value (ADC2), TX PWR (ADC3), PA Temp. (ADC4).
The control lines for DDS1 (AD9951) and DDS2 (Ad9834) are as follows:
The colors are the colors used for the cables in my radio.
The LCD is connected likewise:
The NOKIA5110 LCD has been designed for VDD=3.3V. Please use 10kΩ resistors in the control lines which are not in the schematic! 3.3V are derived more or less closely by switching 2 Si-Diodes in series which results in a voltage drop of about 1.4V. Hence the LCD gets 3.6V DC from the 5V supply chain which is no problem for the module. One big advantage of the Nokia LCD should not be forgotten: It is very quiet and does not produce any discernable digital noise. Thus it is my favourite meanwhile for receivers on the RF bands.
For both DDS modules coupling out the rf is done with symmetrical circuits using trifilar broadband transformers. 10 turns on a FT37-43 core are a good choice. This will enhance gain and reduce spurs.
DDS2 is clocked to 110MHz, but keep in mind, that AD9834 is specified for 75 MHz max. clock rate only. I found out that modules from the “grey market” sometimes fail and produce lousy signals when overclocked. You can see that on a scope when extra peaks appear or with the spectrum analyzer when spurious signal are frequent. I recommend buying with Mouser or anther trusty vendor for example or reduce clock rate in case of problems in signal quality.
Power consumption is not excessive because both DDS modules are for low power application, unlike the AD9850 or AD9835, that draw much higher current. Power rate is 300mA when in receive mode with LCD backlight on.
The C-code for the software has about 2600 lines source code and can be downloaded here.
A standard CB DC supply cable is used here. Unfortunatel the plugs equipped with a cable and fuse holder are widely availabe but the sockets have to be stripped from old CB trasnceivers.
On the air the transceiver performs great. Audio is clear and powerful what the QSO partners often tell me. The receiver is fun to listen to, sounding soft AND precise. Maybe I will do a YouTube video the next weeks to prove it! 😉
All my rigs are for portable, hiking, bicycle trips and travel to foreign countries. I use Aluminum as a basis for the hardware to keep the radio lightweight. With this radio a ground plane made of 0.8mm Aluminum sheet metal has been used that one has been enforced with a lateral additional ground plane carrying the DDS system (see pictures in this article, please). Thus the base frame is pretty rigid and not prone for bending.
Front an rear panel are made from 0,.8mm Al sheets (rear) and 0.5 mm Al sheet (front).
The various subassemblies (DDS, receiver, transmitter) are split into different modules and are seperatelay fixed with bolts and washers mounted to special spacer bolts for screws of 2mm diameter. This ensures better grounding instead of using larger veroboards. Connections are made from flexible stranded hook-up wire and shielded cable for rf and audio signals.
On the undersides of the single boards copper foil is used for lines with GND portential.
Currently I am revising older projects that are in my radio shelf, some of them not finished yet, postponed to a later date, some without a cabinet, some with severe problems with performance and so on. All the stuff that needs a “second chance” ;-). This project is one of this collection. The transmitter did not work correctly (severe parasetic oscillations occurred when the section was driven to power levels >1 watt).
By careful testing and examining I found the reason: The grounding of the rf power amplifier stage was defective due to a connection that had not been soldered properly. After having cured that I found the output was 5 to 6 watts PEP output (very clean). Then, having the project on “GO!”, I finished the design. Thus I got a nice little “vintage style” SSB QRP trsanceiver as a travel or hiking companion:
Frequent readers on my blog know that one thing I really enjoy is building radios based on a minimalist concept. The fewer components you need for a working transceiver, the better it is. At least in my point of view. Here is another one of these “very lean design” transceivers.
The radio originally was designed as a study for my “Old School Transceiver“. After having not built a “real” analog VFO for a number of years I wanted to find out if I still can set up a construction that is really stable concerning frequency. And because it is not very challenging to just watch the result on a frequency counter, a full transceiver had to be built along with the VFO. The VFO was OK, (see later text!) the power transmitter, as mentioned before, was not. Until I had revised it.
The design is another remake of the „Kajman Transceiver“ by SQ7JHM. A design I absolutely love because of its simplicity. The radio basically has been designed for 80 meters (even when lot of websites quote it as a 20m rig) so it shows some weaknesses when adapted to 14MHz without any changes. Thus some improvements had to be made.
Improving performance of the SQ7JHM basic design
Some changes that were top of the agenda to meet my requirements:
The receiver needed a preamplifier for bands where atmospheric noise is not that strong. A dual-gate MOSFET equipped radio frequency preamplifier improves noise figure significantly and can be put into the AGC chain to give more dynamic range and a more pleasant listening experience.
An AGC (automatic gain control) is a good idea if you want to use the receiver in a more comfortable way without the need to lower the volume when strong stations appear. In addition the S-meter reading can be derived from the output of the AGC DC amplifier stage.
A little bit more rf output power can be achieved by using a push-pull amplifier. Linearity also improves to a certain degree when using this design because AB mode combined with separated amplification of the half waves plus suppression of even-order harmonics.
To enhance receiver gain a single stage interfrequency amplifier has been added that is only in use when on receive. It is also connected to the AGC chain.
And, last, a microphone amplifier allows you to talk in a moderate way into the microphone which is good for me because I often have my QSOs when the rest of the family is asleep and not keen on listening to my strange “This is DK7IH/QRP, do you copy?” messages.
The schematic of my enhanced design:
Fascination originates from the fact that you only need a handful of components (OK your hand should not have the size of that of a new born baby!) to set up a working short wave SSB transceiver.
Some thoughts on frequency stability
Careful design is the key for stable operation. This means component selection as well as setting it up on the veroboard.
The basic problem for every conventional free running VFO is temperature and its influence on the size of components. Due to the theory of thermodynamics all materials change their mechanical dimensions with temperature. This is caused by the kinetic energy of the molecules forming the crystals of a solid body. Thermal energy leads to enhanced oscillation of the molecules and therefore the need of larger spaces of each in individual molecule in a crystal. Because we have capacitors in a tuned circuit this will affect the values of all caps (wanted and unwanted ones) to a certain degree.
Something that helps the builder is called “temperature coefficient”. This means that electronic components increase OR decrease their respective value when they get warmer. The first is called “positive temperature coefficient”, the opposite is called “negative temperature coefficient”. So, you might guess, the fine art of radio building involves the knowledge of the characteristic behavior of components when heated.
I quote my findings about temperature behavior listed in the article referred to on the beginning of this text:
Ceramic capacitors: —
Polystyrene capacitor: –
NP0 (C0G) capacitor: no measurable effect
Air coil on polystyrene coil former: +++
Coil wound on T50-6 yellow toroid: +
The more “+” or “-” signs, the more steep the function of T->dC or T->dL is. So you can see: The best choice are polystyrene capacitors combined with coil on a yellow toroid. This combination is likely to outbalance temperature effects. If extra capacity is needed, NP0 caps are recommended.
From the existing principles of building a free running radio frequency oscillator I prefer the Hartley circuit. It uses a tapped coil (tap about 1/5 from the “lower” end) and saves capacitors compared to the Colpitts design. The tap achieves in-phase feedback. The lower you put the tap to the end the lower the amount of fed back energy will be. This leads to more frequency stability because the circuit does not heat up by excessive internal radio frequency. But be sure that oscillation is always strong enough and does not stop. The Hartley circuit is more simple and caps always inherit the risk of thermal problems when poorly selected.
The tuning is done with a Vernier drive and a homemade variable capacitor. For this a foil variable cap of an old AM radio has been dismantled an reassembled with air as dielectric. Lots of experiments were necessary to get the “frequency swing” correct and the basic capacitance to the right area.
Other measures that support frequency stability are :
Low DC power into the oscillator stage (avoids heating the device up by DC current),
Stabilizing voltage for the VFO stage by 2 consecutive steps,
Using a FET instead of bipolar transistor (no PN boundary layers in a FET),
Very loose coupling between oscillator and buffer stage reduce fed back of impedance changes by the output,
Low impedance output with emitter follower,
Avoid metal sheets (spec. Aluminum) close to the tuning elements! Aluminum sheet metal changes its size largely with even low temperature differences.
This oscillator is stable. It needs 5 to 10 minutes to settle which is in the normal range of what can be expected. I then can have it tuned to one frequency and there is a maximum change in frequency < 50Hz for hours. And, to compare with synthesizer technology: NO birdies at all. Really not. I love it! 😉
The mixers and filter section
NE602 and its derivatives have been used in legions of amateur transceivers. Basically designed for cell phones and small cordless phones radio amateurs quickly have found out that this mixer IC can be the universal mixer in lots of possible amateur radio designs. The main weakness is its low IMD3. But for a 14MHz rig the risk of appearance of strong out-of-band signals is not that likely. Besides, the selectivity of the receiver’s input section supports this. Strong in-band signals did not appear so far due to low band conditions. We’ll have to see how the receiver performs here.
On the other hand NE602 gives a good sensitivity which makes it ideal for radios on the higher bands where signal levels are not so high.
The NE602 has a balanced input AND a balanced output. This allows the designer to get two different signal sources to the input then subsequently mixed with the oscillator signal. As well the two outputs can be used to send the mixed signal to different paths.
This is what is the basic idea behind the design described here.
The mixer that is used together with the microphone to produce the DSB signal by mixing the audio signal with the local oscillator (LO) also serves as the product detector on receive by mixing the interfrequency with the LO. Correct signal path is set with the two relays depending on the fact you are either on transmit or receive mode.
The same principle is for the other mixer. It is transmit mixer or receive mixer, depending on the position of the relays.
The relays connect the SSB filter either to the input or the output of a distinct mixer. A graphical presentation should make it clear:
RX amp and interfrequency amplifier
These 2 stages are more or less the same. They provide 2 to 12 dB of gain depending on the AGC voltage applied to gate 2 of the dual gate MOSFET. In this version of the radio a potentiometer of 20kΩ is used to have the possibility to lower the DC voltage manually, by doing this an MGC (manual gain control) is achieved in a simple way.
A bipolar transistor and the inevitable LM386 amplify the filtered audio signal from the product detector to a volume that can be discerned even in a louder environment. The audio low pass filter prior to the AF preamp should be selected due to the users individual preferences concerning tone pitch of the audio signal.
RF power amplifier
This is more or less my standard power amplifier for small QRP rigs. I put stress on linear amplification, so I use emitter degeneration and negative feedback in collector circuit to get best IMD3 results. Even if the circuit could deliver one or two more watts I let the output power level at about 5 watts pep.
Here ist the result of a dual tone modulation:
Voltage division is 10 volts per cm, so this is 45Vpp which equals to about 5 watts max. peak output. Quite OK for QRP. And here is the spectrum of a 2-tone-modulated signal:
The whole transceiver is built on a 12×8 cm Veroboard (4.7″ x 3.1″). There is only one layer. The cabinet is 4 cm high (1.55″), 14 cm long (5.5″) and 9 cm wide (3.5″).
Left the vernier drive with the homemade capacitor attached. Left of the 9MHz filter you can see the LO, more far left the S-meter (from an old CB radio) hiding the audio amps. The 2 mixer ICs and the relays are sited around the SSB-filter. On the right side the power amp partly hidden by the DC switching board.
Well, that’s the story how a nearly failed project was saved from the scrapyard and came to life by carefully searching the faulty element in the circuit.
This article describes the “Cigarette Pack” SSB QRP transceiver” for 14MHz that I first had mentioned some months before. Recently, when taking it from the shelf, the transceiver dropped to the floor and was severely damaged. This lead to serious defects in the front panel area, the main frame, the cabinet and so on. The interior parts were, luckily, not affected by the crash. So, I had to revise the whole radio, make a new front panel and cabinet, ply the frame straightly (as far as possible) and so on. This is the full description of the rig now to complete the files here. The good news: The radio is fine again and fully operational! And the even better news: I still have not started smoking!
During reconstruction the transceiver has been extended for about 5 mm so that overall length now is 100mm (3.9 inch). This was done because I intended to build in a loudspeaker. The other dimensions remain unchanged: Width is 52mm (2 inch.), height is 30mm (1.2inch). OK it is slightly longer now than a standard pack of cancer sticks, but who cares? Total cabinet volume is 150cm³.
The transceiver is based on the “Micro 23” rig, that I have described here. Some simplifications of that already simplified radio have been made. Here is the full schematic of this even smaller transceiver:
Very simple rigs like this one always use parts of the circuit for receive and transmit purpose. Here these parts are the 2 mixers (NE602), the SSB-filter and the interfrequency amplifier.
Signal flow schematic
The NE602 has a balanced output. With mixer 1 only one of them is used. If higher gain is desired, a broadband (or even better a tuned LC circuit) transformer could be used to connect pin 4 and 5 (the mixer outputs) in push-pull mode. I did not do that to save the transformer.
The signal flow can be derived from the design:
Receive mode signal flow
From the antenna relay (not drawn) the rf energy runs through a 2 pole LC filter for 14 MHz. The coils are wound small TOKO coil formers, all respective data is given in the schematic. Coupling is loose via a 3.3pF cap.
NExt stage is an rf preamp for 14MHz with a broadband output. The acitve element here is a dual-gate MOSFET.
After having left this stage the 14MHz signal travels through another 470pF capacitor. This one has high resistance for audio frequency and low for rf frequencies due to the equation: XC =1/(2*PI*f*C). The signal is then fed, together with the audio signal from the microphone (when on transmit), into mixer 1 input on pin 1. The 1k resistor prevents the rf energy from flowing into the microphone circuit. The two signals are separated from each other by simply exploiting reactance and resistance in a rather clever way 😉.
When receiving the Si5351A clock chip is programmed in a way that the VFO signal (23 MHz) is present on output CLK0. It is fed into mixer 1 via a small cap to prevent overloading of the mixer. The Si5351A breakout board delivers about 3 Vpp. clock signal, so this must be reduced to about 200mVpp. A 5.6pF capacitor is OK here.
The resulting signal is sent to the SSB filter (a 9MXF24D) that is terminated with 1kOhm and 20pF in parallel. The wanted SSB signal is present at the output of the filter.
Next stage is the interfrequency amplifier, equipped with a dual-gate-MOSFET semiconductor. This one is connected to the AGC chain, on receive a variable voltage is applied to gate 2 (range 0 to 6 V), on transmit the AGC is fully powered to ensure maximum gain.
Next is mixer 2 which is the product detector when receiving. The signal (9MHz +/- sideband shift) is applied to pin 6. Due to the fact that this mixer also serves as transmit mixer, the two signals are taken from the two mixer outputs on pin 4 (serving as audio output) and pin 5 (serving as rf output for transmitting).
Two audio amplifiers (preamplifier and power stage) give a sufficient signal level for an 8 ohm loudspeaker or a headphone.
For the loudspeaker I tried out the tiny ones for smartphones with good success. Only the volume was a little bit low. Then I found another speaker in an old toy of my daughter that turned out to be very much OK for this transceiver. Its diameter is about 3 cm (1.2 inch) and just fits in the housing.
Transmit signal flow
The microphone in this radio is an electret one. The advantage is that these microphones have an internal preamplifier equipped with a field-effect-transistor. The output voltage is fairly high, about 1Vpp. when normally speaking into it. Therefore an audio preamp is obsolete. The microphone signal is directly fed into pin 1 of the first mixer. On transmit the Si5351 signal generator is switched that the 9MHz (+/- sideband shift) signal is fed into pin 6. The SSB filter eliminates the unwanted sideband, the interfrequency amplifier lifts the SSB signal to an appropriate level. The TX mixer is fed with the 23MHz signal resulting in a 14 and 37 MHz signal. The TX band pass filter cleans the signal from the unwanted 37MHz component resulting from the mixer process.
RF power amplifier
The power amplifier is a 3 stage circuit. Stage 1 (preamplifier) brings the signal to about 10 mW. This is coupled into the driver stage via a cap of 0.1uF without any further impedance matching.
The subsequent driver stage shifts the signal level to about 200mW. Linear amplification is ensured her (as well as in the previous stage) by negative feedback in the collector circuit and emitter degeneration with a non-bypassed resistor to GND. An output transformer (winding rate 4:1, impedance rate thus 16:1) lowers the impedance of some 100 ohms to a few 10 ohms present on the input of the final amplifier stage.
The final amplifier brings up a signal level of 3 to 4 Watts PEP. This stage is in AB mode, the appropriate bias is achieved by the 1k resistor going to +12V TX and the current to GND via the silicon diode. This diode must be thermally connectod to the final transistor to stabilize the bias.When the transistor heats up, the silicon diode increases the current through it thus decreasing bias to the transistor.
The 68 ohm resistors serves 2 purposes: First it prevents the input signal from being shorted by the bypass caps in the bias circuit and it stabilizes the rf behavior of the stage by limiting the gain because certain amounts of the input power are led to GND. This prevents self-oscillation.
DC ad the collector is fed through a radio frequency choke to hinder rf from flowing into the DC line. Radio frequency is directly fed into the low-pass-filter. The output impedance of this stage is roughly 50 Ohms, so the filter can be a 50 ohm circuit with a cutoff frequency slightly above 14MHz.
The VFO section
The Si5351A clock chip used here has three frequency outputs that can be set individually. Only CLK0 and CLK1 are used in this radio. The Si5351A chip is programmed by software in the following manner:
Receive: CLK0 is the VFO, CLK1 is the BFO.
Transmit: CLK0 is the BFO, CLK1 is the VFO.
The microcontroller reads the tx/rx status and switches the frequencies respectively.
The radio is a full SMD design on a 0.1″ pitch double sided Veroboard:
The control panel on the left with tuning knob and volume set. The 64×32 pixel OLED between these controls. Following the microcontroller behind the fron panel (here covered). The controller is an ATmega168 on an Arduino Pro mini board.
The isolated board left of the SSB is the AGC section. The receiver and transmitter shared parts follow, the TX band pass filter is in the foreground. The power transmitter is on the right behind the shield. The shield is necessary to avoid unwanted oscillations when rf is coming back from the power transmitter to the band pass filter prior to the tx section.
On the right there is the SMA socket for connecting the antenna plus a 3 pin header for connecting a headphone. When there is no headphone in use a jumper connects the internal speaker to the speaker line. VDD is applied via a standard DC connector.
The underside of the board has only some SMD components and the wiring on it:
“On the air”
My longest distance achieved with this transceiver (after rebuilding it) has been R2DLS near Moscow who gave me a “59”-report. The antenna in use is, as always, a Deltaloop.
In this paper we will discuss a single sideband amateur radio transmitter/receiver for the 40 meter band that has been designed to ensure good performance characteristics with reasonable number of parts (no “overkill” in component use), particularly concerning the receiver. Circuit simplicity and over-average performance were to be combined.
The background: Some years ago I had built the ancestor of this transceiver and afterwards posted an incomplete series of articles (starting here). The transmitter was considered to be quite OK (I could even work a station from South Korea when operating as GJ/DK7IH some years ago) but the receiver was weak.
The shortcomings originated from the rf preamplifier I used together with the 1st mixer, an NE602. The latter had severe problems to cope with the high signal levels on the 40 meter band from out-of-band broadcast stations transmitting on the 41m band (f>7200kHz) or from very strong amateur stations transmitting in-band. This is caused by the technical specs of this Gilbert cell mixer. NE602 has been designed for mobile phone applications and not for shortwave radios. Its IMD 3 is only -15dBm whereas it is able to detect weak signals (-119dBm with an S/N ratio of 12 dB) according to datasheet. Due to this NE602 was excluded from being used at least in the receiver.
Another point was that the rig was too small and too densely packed to be called “service friendly”. Thus I dismantled the radio some times afterwards and had in mind rebuilding it with another receiver and a little bit more space inside.
The project has had to meet certain requirements that I would like to point out first:
Frequency generation: Dual-DDS-System: AD9835 as local oscillator and AD9834 as VFO. ATMega644A as MCU (Download source code here)
Receiver: Single conversion superhet, 9 MHz interfrequency with commercial filter (supplied by http://box73.de) shared by transmitter and receiver and relay switched, “NE 602-free zone” ;-), 4 dual gate MOSFETs in rf preamp, rx mixer, if amplifier and product detector, audio stages with BC547 as preamp and LM386 as main audio amplifier.
Edit: I found that there was strong signal of self-reception around 7.100kHz which was not a spurious signal from one of the DDS. It has been a mixing product of one or two oscillators together with a signal from the microcontroller. So I changed the interfrequency to 10.7MHz which cured the problem. I tried to calculate the issue but was not succcesful because I do not know all the frequencies in the microcontroller. I think it is most probable that it is a harmonic of the PWM signal I use for controlling the LED front lights.
Transmitter: 4 stages, 3 of them in push-pull mode, Siemens made mixer IC S042P (really old fashioned, but still available) as DSB generator and TX mixer, rf amplifiers (2N2219A) after filter and tx mixer.
Design: Really “cool” with blue backlight. Sandwich built, not the size of a “micro transceiver”, but handy for travelling.
The Block Diagram
The diagram can be derived from the old project, it is nearly the same:
The basic outline of the radio is standard and should not be further discussed.
Dual DDS (VFO and Local Oscillator (LO))
This time I wanted to use 2 digital oscillators. The reason was just to have fun. 😉 Here is the schematic:
The source code has got about 2200 lines. With the GNU C compiler this leads to a HEX-file of about 43kB. Because of this the controller had to have a little bit of more memory. A “644” is a good choice here. It is clocked internally to 8 MHz clock rate. Radio and user data (user operated keys, S-Meter, TX PWR meter, temperature sensors attached to final transistors) is lead to the analog-digital-converter (ADC) of the MCU. Rotary encoder (optical) is fed into digital inputs. Integration of an RTC is projected but not done yet.
Here an AD9834 is used. It is overclocked with 110MHz clock rate. For my receiver with a DDS chip purchased from Mouser this works without any abnormality. With a a chip from the “free market” (ebay) I found that there were strange clicks in the signal. So, I do not really recommend overclocking under any circumstance and/or not to such a high degree.
This DDS is is not terminated with a low pass filter. Due to the high clock rate there is no clock oscillator feedthrough which is supported by the design of the following amplifier having an audio frequency transistor in the last stage (BC547 and later BCY59) that limits high frequency components due to its early gain decay in the frequency spectrum. The two stage amplifier has been designed for excellent linearity to prevent impurities in output spectrum.
The first peak showing the 16MHz signal and the next peak is the first harmonic about 30dB below. Other peaks are from local sources (PC, Printer).
The sine wave also looks quite OK:
This one contains an AD9835 synthesizer clocked to 50 MHz. An LPF here is mandatory. A simple but linear amplifier brings the signal up to 3Vpp which is OK for driving the dual gate MOSFET in the receiver. For the transmitter mixers this amount of voltage is too high, small capacitors reduce the voltage to an acceptable value.
From another project that I once had built and that is not more in use, a dive computer, I had a 4 lines/20 characters text display that is fairly large. This was to be designated as the LCD for this transceiver.
Building a receiver for the 7MHz amateur band is challenging. On one hand the circuit should be very sensitive for weak signal reception, particularly during day when the band conditions are low due to solar radiation and density of the D-layer. This means the receiver should have a higher gain whereas noise figure does not play a predominant role due to band characteristics with high atmospheric noise on 7MHz.
Next request is high dynamic range to eliminate the spurious signals that occur when front end stages are loaded with high input signal levels.
And last but not least AGC control range should be as wide as possible to cope with weak and very strong signals without the request to intervene by adapting manual gain control. For this a preamp also benefits.
Active mixers like the NE602 show low performance under these conditions. Some high-current mixers like the SL6440 exist, but there are alternatives. On one hand the classical diode ring mixer might come into perspective, otherwise Dual-Gate MOSFETs are well known as having a fairly good ability to cope with high signal levels and so don’t tend to deteriorating the receiver’s performance severely. Besides they offer some gain and low noise figure (which has not been the main objective in this case) and the circuit is very compact and therefore it was the best choice for a receiver that had been intended to be constructed onto a board of 6 x 8 centimeters.
After these thoughts the following circuit turned out to be the right onset for a receiver inside the projected rig.
Circuit explanation (Receiver)
On the left we start with a 2 pole LC band pass filter for 7 MHz. The coils are wound on TOKO style coil formers (5.5mm size), winding data and parallel capacitors are given in the drawing. The coupling capacitor (2.7pF) between the two LC circuits is very small for such a low frequency. This makes the filter response curve sharper but leads to a slight weakening of the signal coming through the filter. But as the whole receiver has plenty of gain and a very good noise figure, this is the reason why some weakening of the input signal is acceptable.
Next is the preamplifier for the received band. It is connected to the AGC chain. You can expect some 25 to 30dB gain swing by driving up gate 2 of the dual gate MOSFET from 0 V to 6V. A 1:1 voltage divider decrease the 0..12V AGC voltage to 0..6 V where th3N205 MOSFET is close to amplify with maximum gain. Exceeding 6 to 7 volts does not result in significant more gain swing, so I usually drive the MOSFET from 0 to 6.5 volts UG2 (with 13 Volts of supplied voltage.
The coupling when going from the preamplifier to the receiver mixer is in broadband style. The 3N205 has a very high gain and tends to self-oscillate. A second LC circuit makes the device more prone to going self-resonant and hence produce unwanted signals.
This mixer is very simple and needs only a few components. Both signals are fed into the gates of the dual gate MOSFET. Rf goes to gate 1 whereas gate 2 (the AGC input) is fed with the oscillator signal). Gate voltage depends on the voltage drop at the source resistor and therefore is stabilized. The oscillator signal should be in the range of 2 to 3 volts rf (pp) for a dual gate MOSFET. Lower values will deteriorate the performance of the mixer, e. g. its dynamic range. This signal switches the semiconductor and a superposition of the two signals occurs thus leading to the production of sum and difference of the original frequencies. These signals are fed into…
The SSB filter
which is a commercial one (Supplier box.73.de). The reason why I don’t ladder filters anymore is that I found it extremely difficult (not to say impossible) to get a symmetric filter response curve thus making the lower and upper sideband of the receiver sounding different even when the carrier frequency has been adjusted very thoroughly.
The filter is used for the SSB transmitter as well. To ensure maximum signal separation between the two branches (tx and rx) and between filter input and output I again us a high quality rf relay made by Teledyne. When choosing a relay intercontact capacitance is crucial. It should (if possible) be < 1 pF.
Don’t forget a clamp diode to VDD over the relay coil to eliminate high voltage voltage peaks generated by self inductance when the coil is switched off. Voltages up to 100 Volts can occur. This might damage the transmit-receive section of this transceiver that is equipped with semiconductors only and does not use a relay.
This circuit is the same like that of the rf preamp. It also is part of the AGC chain, thus delivering another 25 to 30 dBs of gain swing so that overall gain swing is around 50 to 60dB. In practical research over a long period of observation I found that with an antenna delivering high signal voltage (Delta loop) it was not possible to overdrive the receiver to a level where signal distortion was audible.
A tuned circuit is also placed here to increase gain. Tuned amplifiers usually have higher gain than broadband ones. It is highly recommended to ground the metal cans of the coil to prevent any self-oscillation. But as I found out, this amplifier is not very prone to go to self-oscillation state.
Here again a dual gate MOSFET is used. The circuit is nearly the same like the RX mixer except from the output section. We can see a low pass filter here, consisting of 2 Cs (0.1uF) and a resistor (1k). You can use a radio frequency choke instead, 1mH is recommended.
This section consists of two parts, a preamp (with bipolar BC547) and a final amplifier (LM386 IC). It is well-known that this IC tends to oscillate. One measure to prevent this is to keep leads short, switch a low-pass filter (capacitor 100uF and R=33Ω) into the VDD line and to reduce the gain capacitor between pins 1 and 8 to a degree where self-oscillations terminate.
A switching transistor cuts off the audio line by short circuiting it when on transmit. This eliminates any noise when switching. The rx/tx switch now is 100% “click free”. A very pleasant way of operation. 😉
This is another re-use of a circuit I have frequently used before. It is desired to reduce its output voltage down to 0 volts when a more or less strong af signals appear at the input. The agc voltage is derived from the audio signal of the receiver. Some say that this is not the best choice because you need more time (an af cycle last much longer as an rf cycle) for the waveform to generate the regulating DC voltage.
Nonetheless I have never observed popping or unpleasant noise from incoming very strong signals. The agc response rate is so fast that you won’t notice that it just has regulated even when a strong signal comes in. Only with very, very strong signals a slight “plopp” sound is observable but it is not unpleasant.
A second capacitor can be switched in parallel to the 33uF one. This can either be done by a transistor switch (like shown in the schematic) that in this case is controlled by an output PIN of the MCU. An alternative that I found later is to use the MCU pin directly to switch the cap. When not using the additional cap you must switch the pin as an input so that there is no positive voltage from the pin to the circuit. When you intend to ground the transistor (agc in “slow” position) then the pin mus be set as output by defining the DDR-register respectively AND the pin must be set to 0. So you can get rid of the switching transistor.
Another possibility would be to derive the agc from the interfrequency signal. The problem that occurs in this case is that you have to decouple the local oscillator (bfo) very carefully from the place where agc circuit is placed. Otherwise you are at risk to detect the bfo signal by the agc which leads to reduced response range in the agc. In addition this receiver uses a higher rf voltage level for the mixers (2 to 3 Vpp each). By this the amount of stray energy is higher inside the circuit and thus this rf energy might be detected very early by the agc.
In the emitter line there is a resistor (68Ω) which produces a voltage drop when the transistor is driven. This is fed into the ADC of the microcontroller driving the S-meter display part.
First the circuit:
This amplifier is a simple common-emitter circuit with the directly grounded emitter of the BC547 transistor. This circuit is linear only for low input voltages but suitable for the connected dynamic microphone since this does not produce more than some millivolts of audio energy. Bias comes from the 390kΩ resistor. At the input you find a 2.2nF capacitor from base to GND which helps to prevent coupling in rf energy from the transmitter to the audio stage and thus leading to an impure signal.
The DSB generator + amplifier
The amplified microphone signal is used to produce a double-sideband signal. The ic I use here is an antique but still available part by German manufacturer Siemens, the S042P. It includes a so-called “Gilbert-cell” mixer and an oscillator but the latter is not used here (DatasheetApplication note (in German)).
The S042P mixer needs some more components compared to the well-known NE602 integrated circuit but fewer ones than the MC1496. It is designed for 12V usage, thus no voltage regulation is required.The ic can be applied in balanced mode or non-symmetrical. To save components I use the unbalanced circuit alternative. A slight loss in output power is acceptable in this case, there are amplifiers post each mixer in this transmitter.
Ic gain is about 16.5 dB, DC current is about 3 mA.
A crucial point is the signal level of the local oscillator. S042P needs only some hundred millivolts of oscillator voltage. To prevent overdriving I experimented with different values of the coupling capacitor. 5.6pF seemed best because the LO produces some volts peak-to-peak.
Following there is an amplifier that is a standard circuit and has been tuned for maximum linearity in order to reduce distortion to a minimum (which is also true for the following stages). You can see the well understood 2 master ways of achieving max. linearity in an amplifier stage:
Negative feedback between collector and base (i)
Emmitter degeneration (II)
i) The first measure goes along with the 2.7kΩ resistor between collector and base of the transistor. This resistor provides positive dc bias voltage to the base and leads 90° out-of-phase ac voltage to the transistor’s input. This reduces gain and therefore distortion. But due to the fact that the whole transmitter strip has plenty of gain, this loss in gain is not a serious problem.
ii) The 10Ω resistor in the emmitter line is not bypassed by a capacitor. This stabilizes the circuit. When the current through transistor increases the emmitter voltage will rise (according to Ohm’s law) and the voltage between collector and emmitter drops. This reduces voltage difference between base and emmitter and hence also reduces gain.
The coupling to the next stage is done by a capacitor of 0.1uF. This causes some impedance mismatch. But that is as well not a big problem because the gain reduction here helps to prevent the whole transmitter from unwanted oscillations by diminishing overall gain.
Here the second S042P is used. The 9 MHz SSB signal is coupled to pin 13 of the ic, a DC connection is established to pin 11. These two pins represent the base connectors for the two current control transistors and should be bridged by a DC resistor in this circuit.
The 150Ω resistor from pin 10 and pin 12 to GND defines the gain of the mixer. Here you can use down to 150Ω but should have a resistor towards VDD to limit current and avoid excessive heating of the device. In this case another 150Ω is used.
VFO signal is coupled symmetrically to pins 7 and 8 via a small trifilar toroid. See schematic for details and please note that center tap is not used here. This is in contrast to the output transformer where the tap is used to feed supply voltage into the mixer.
Another 7 MHz band pass filter terminates the mixer, data for coils and capacitors is in the schematic.
This amplifier has got 4 stages and except from the first one all are in push-pull mode. The power distribution for these 4 stages is as follows:
The first of the 4 power stages is the same as the post dsb generator amplifier so there is not more to add concerning this stage. Rf energy is taken out via a transformer with a primary and a tapped secondary winding. This is to provide the balanced structure necessary for the following push-pull stage.
This is the first push-pull stage. Its bias is derived from a voltage divider connected to the tap of the input transformer.
Please note: In contrary to the schematic I have installed 2 devices of the 2SC1973 type because the signal turned out to be much purer with these ones on the spectrum analyzer.
A tapped output transformer feeds the amplified rf energy to next board. Output impedance is 50Ω. The coupling to next stage then is done via a shielded cable of (nearly) the same impedance.
This one has an input transformer also center tapped. The tap goes to a bias network consisting of a current limiting resistor (1kΩ), two diodes forming the lower part of a voltage divider and some capacitors as part of a low pass filter to avoid coupling in of radio frequency (rf) energy. The two diodes must be thermally connected to the cases of the transistors. In case these heat up, the diode increases its conductivity thus reducing its resistance. The bias voltage drops and heating is stopped. So, thermal runaway is prevented.
For these two stages (predriver and driver) DC is fed through low pass filter (RFC and 2 caps 0.1uF) to prevent coupling of rf energy via the VDD line.
This stage receives input from a balanced structure without a center fed transformer. Instead bias current is linked in via a network of radio frequency chokes and two resistors of 5.1Ω each.
Bias is provided by a current regulating transistor and should be set to about 100mA.
The MRF455 transistors are mounted directly to the aluminium structure of the sheet metal carrying the whole transceiver boards. When mounting them to the Veroboard I did not solder them directly. I used 1.6mm screws and washers to press the brass connectors to the copper strips of the amplifier board:
With this I could have been able to remove the precious transistors without having to unsolder them when the device might have turned out to be a failure. But it was not, thank God!
The output transformer is the one I have used in my old 14MHz PA and the ancestor of this radio. It is from an old ATLAS 215 transceiver and I hope that this will be the final place for the transformer.
Two temperature sensors (KTY-81-210) have been installed to measure the temperature of each transistor. They are connected to the microcontroller via voltage dividers (see schematic, please!)
Low Pass Filter and Power Measurement Unit
For the low pass filter I use 2 toroids T50-2. These might appear small but from one source (that I have forgotten) I remember to have found that for 50 watts of power this core is still suffice. Metal powder cores can stand much more power compared with same sized ferrite toroids.
The power measurement unit consists of a network that starts with a resistor of 12kΩ to ensure a significant voltage drop in signal level, then two rectifier diodes (1N1418 or equivalent) follow, some low pass filtering eliminating the last rf energy and the resulting direct current voltage is fed to a variable resistor to set an adequate voltage level for the ADC in the microcontroller.
The rf output made out of a two-tone audio signal measured at the antenna connector:
The spectroscopical analysis shows the signal on the f -> V figure:
A very simple circuit. Two PNP power transistors are used but they don’t have that much to do. They are only designed for switching the low-power parts of the radio. The high current to the drivers and final amplifiers is permanently present in the collector lines but the bias lines are tx/rx-switched and go to 0V during receive periods. This reduces requirements for the power rating of the switch board.
When pushing the PTT the base of the lower transistor is pulled to GND. So it becomes conductive and TX DC is applied. Via the diode the upper transistor loses its negative voltage and becomes non-conductive.
One interesting thing was the blue backlight to illuminate the front panel controls. It is made using SMD LEDs that are soldered to small pieces of Veroboard and fixed with 2-component glue to transparent light-scattering plastic bought from a local supplier for architects and designers. This material is used for making models of houses and stuff like that. As light distributor this material is excellent. The LEDs are powered by a linear transistor connected to the pulse width modulation (PWM) output of the microcontroller so that light intensity is adjustable.
Hint: When programming the PWM functions it might occur that PWM frequency is audible in the receiver. If something like that occurs another frequency can be selected without changing the performance as soon as it is high enough that human eyes aren’t able to recognize a flickering.
The covers used for the labels and the LCD shield are made from 2mm acrylic and fixed with screws of 1.6 respective 2mm diameter.
The two push-buttons right in top position consist of two bars of acrylic (4.2mm diameter) and are having mechanical contact to small spring-loaded switches behind the front panel:
Directly under these acrylic bars there are two LEDs shining into these rods and because of total reflection inside the tubing the optic conductor is sending the light to the front side when the LEDs are powered on. That is how it looks at night:
This is a sandwich construction again. On the first side there is the DDS board (left), the receiver (center) TX mixer and preamplifier (right) and the SSB generator (back). Also there is a 5 lead connector holding the 5 ISP lines (MOSI, MISO, CLK, RESET and GND). This makes firmware updates easy because you don’t have to open the case when you want to update software.
The other side holds the TX low pass filter plus power measurement unit (left), the power amplifier (center) and the predriver and driver (right). In the back you can see the rx/tx switch board:
“On the air”
Again big fun this transceiver! During the ARRL DX contest last weekend I could work some statesiders. With Delta Loop and 50 watts, fairly OK. Working Europe all day is no problem with 50 watts.
During the first QSOs I had reports that the audio sounded clear but somehow “narrow”. I had used an electret mike that time and could not use a dynamic one because the preamplifier following the microphone did not have enough gain. Then, to solve this problem, I decided to do a full reconstruction of the SSB generator board. The one then had used had an AN612 mixer integrated circuit (from an old CB radio). This one was dismantled and replaced by the S042P board. The change took me 3 hours to develop and solder but it paid. I use a Motorola dynamic microphone now that has a very rich and clean sound. I monitored it on a web based SDR receiver, made a recording and found it to be OK.
OK, dear fellow hams, that’s the story so far, some supplements will sure be made, so stay tuned!
Having deferred the work on the “micro multibander” for some time I finished another small QRP rig (this one for 7MHz) that is suitable for my summer excursions by bike or hiking the local mountains here in the State of Rhineland-Palatinate or the Black Forest that is not that far away on the other side of the Rhine valley.
Besides, this transceiver to be discussed here is some sort of a “remake” of a 20 meter rig I built 3 years before. And this time, the transceiver really fits into a shirt pocket without having to wear “XXXXL”- clothing. ;-):
General circuit description (instead of presenting a block diagram)
The rig uses two mixers NE602 plus one filter as central elements. The signal way is reversed when switching from receive to trasmit mode. This is done by 2 relays and is a well known technique for simple QRP rigs. You will find lots of equivalent ideas on the internet (Example 1, Example 2).
But not to ignore the shortcomings of these designs: They are somehow inferior to my requirements, particularly concercing receiver performance. I prefer to have higher signal gain and an AGC circuit. AGC for me is a must. But these designs can be expanded easily, so I added an AGC controlled interfrequency amplifier with dual gate MOSFET BF998 into the receiver’s signal path enhancing performance significantly.
The frequency generation of the superhet transceiver scheme is simple: Again I use one interfrequency (i. e. 9MHz). The VFO is DDS based on AD9835 operating below the desired radio frequency, which means that it is set to the range of about 2 MHz. Due to this low frequency you could replace the DDS by a VFO if you don’t like the relatively complex work with the software programming and microcontroller stuff). A 2MHz VFO can also be made very stable, so this is an alternative not to be ignoered.
Due to the fact that the schematic is not very difficult to analyze you are kindly requested to refer to it for further talking:
In the center of the schematic you can see the main elements of the circuit: One SSB filter (9MHz), correctly terminated by 2 resistors of 1k each (to ensure proper filter response curve) and two relays with a double set of switches. These relays reverse the way the signal travels through the filter. The advantage of this: You can use the integrated oscillator of the NE612 controlled by a crystal and a tuning capacitor to set the carrier frequency correctly for the lower sideband because the mixer is used as SSB generator and as product detector in common.
A word on chosing the proper relays: An intense examination of the relays’ data sheet is essential. I built a prototype of this transceiver on a breadboard prior to soldering the components to a veroboard. I found that some SMD relays have signifikant coupling capacities between the unused relay contacts (in the range of some Picofarads). So stray coupling was a severe problem. Later I used some second-hand Teledyne RF relays that I had purchased via ebay two years ago (price originally 50€!) for 1€ each. These relays are absolutely superb!
Before we go: In the circuit scheme above I missed out the antenna switch relay because I think every homebrewer knows what to do in this case. 😉 So the receiver’s signal path starts with a band filter for 7MHz consisting of to tuned LC circuits. The coupling is relatively loose. As coils I use the well known coil formers in TOKO style with 5.5mm outside measure.
Coil data for the 7MHz band pass filter (BPF) is 39 turns primary and 9 turns secondary of 0.1 mm enameled wire. The respective capacitor is 33pF. This is a high L to C ratio which gives you excellent LC quality factor. This is mandatory especially when working on the 40 meter band, because of the strong broadcasters starting from 7.200 kHz intermodulation might be a problem when the receiver is connected to a high gain antenna and broadcasters’ signals might overload the first mixer (remember that NE612 has a relatively low IM3!). If you still should have problems coping with too strong out-of-band signals you can reduce the coupler from 4.7pF down to 2.7pF.
In practical terms I could not detect any unwanted signal products even when using an antenna with high rf output voltage. One reasons for this is, that there is no rf preamplifier for the receiver. This avoids overloading the first mixer generally.
The NE612 has two mixer inputs and two outputs. This makes it very suitable for this sort of radio. In receive mode pin 2 of the right NE612 is used as signal input. VFO signal is fed into pin 6. The resulting mixer products are taken out from pin 4. Next the 9MHz filter follows from right to left.
The 9MHz IF signal then is fed into an IF amplifier. This one is equipped with a dual gate MOSFET (BF998), gain is about 15dB when full AGC voltage is applied wich leads to about 6V by the 1:1 volatge divider in the applied to gate 2 of the MOSFET.
The left NE612 is the product detector. I use the internal oscillator with a 9MHz crsytal and a tuning capacitor here. This saves building an extra oscillator and simplifies the rig again.
One AF low pass filter made of 1k resistor, 100uF rf choke and a 0.1 uF capacitor eliminates high frequency remainders generated by the mixing process.
The audio stages are also made simple: One preamplifier (using bipolar transistor in grounded emmitter circuit) and a final stage with LM386 transform the signal to a level that is sufficient to be fed into a small 8 ohm loudspeaker or a set of standrd MP3-player headphones. Because the rig is very small and there was definetely no space for a loudspeaker I use headphones instead.
Keep an eye on the power supply switching of the two audio stages. The problem was to eliminate the switching click and pops to a minimum and to avoid acoustic feedback when unsing a loudspeaker. So the audio preamp is only connected to DC on receive. When switching to transmit the charged capacitors avoid instant cut off supplying some Milliseconds DC to the amp until significantly discharged. The main amplfier on the other hand is connected to permanent DC supply. So it won’t pop when switching from tx to rx an vice versa but can cause feedback. To avoid feedback a transistor is used to cut the speaker/earphone from the power amplifier.
AGC is audio derived. A two stage amplifier provides a DC voltage analog to the audio input sginal strength. First amplifier stage is a common emitter bipolar transistor supplying sufficient audio voltage. This voltage is rectified by a two diode circuit letting only the positive halfways pass. You can use silicon diodes (1N1418) oder Schottky diodes here. An electrolytic capacitor (100uF/6V) provides the time constant respectively DC decay once the signal has disappeared. Output of the DC stage is split. The collector is connected to 12V via a 4.7k resistors causing a voltage drop when the transitor’s conductivity increases. The emitter is fed to the ADC of the microcontroller (pin ADC1) causing a proportional voltage to the voltage of the applied audio signal so that on the OLED an S-meter can be displayed.
An electret microphone picks the operator’s voice. The signal output level of these microphones is high enough to drive the left NE612 (which serves as balanced modulator in this case) directly. Signal input for the mixer should be 200mV RMS according to data sheet. An electret produces about 0.5 to 1 V pp if spoken with a decent voice in the distance of some centimeters. So you have more than enough audio signal power for the modulator.
BTW: Carrier suppression of the modulator is excellent. I achieved 56dB without doing anything else!
The resulting DSB signal then is fed into the SSB filter, the SSB signal subsequently is directly sent into the right NE612. A band pass filter for 7 MHz eliminates the unwanted mixer products. You should have 400 to 500 mV pp of rf signal here when the transmitter input is fully driven. I recommend a two-tone test generator to check out the linearity of this and the remaining amplifier stages!
Next parts of the transmitter are a band pass filter (same coils and capacitors like th rx bandpass filter), a preamplifier and a driver. The later should put out about 150 mW into a 50 ohm load. They are made more linear by emitter degeneration (4.7 and 2.2 ohm resistors for predriver and driver) and negative feedback. This helps to ensure that transmitter performance is fine when IMD3 products are concerned even if the main IMD3 problems usually occur in the final stage.
To transfer the rf power into the final stage proper impedance matching is mandatory. Input impedance of the final stage is fairly low (<10ohms), therefore a broadband (down)transformer is used. Data is: Core T37-43, primary 12 turns, secondary 4 turns of 0.4 mm enamled wire.
Last stage is a single ended linear amplifier in AB mode equipped with a 2SC1969 rf power transistor by eleflow.com.
BIAS circuit: The combination of the 1k resistor, a silicon diode (1N4002 or equ.) and a capacitor sets up the correct bias. Bias is fed into the cold end of the input transformer. Quiescant current should be around 40mA. A good thermal contact between the diode and the transistor is recommended. As the transistor gets warmer the diode will increase its conductivity so reducing bias current. This will prevent thermal runaway effectively!
To avoid bulky output transformers the PI-filter (7MHz LPF) is part of the tank circuit of the final amplifier transistor. For this power level this is an acceptable and practical solution because the output impedance of the stage is nearly equivalent to 50 Ohms. A certain mismatch is not a severe problem. DC to the final transistor is applied via an rf choke, for exact data please refer to the schematic!
T2 helps to suppress unwanted signals that I encountered when taking the transmitter from the dummy load test environment to a real antenna. I observed unwanted parasetic oscillation in the range of about 1MHz. T2 has a low reactance for this frequency range thus eliminating the oscillations in a reilable way by short circuiting them towards ground.
Powered with 12.5V DC the transmitter will put out slightly more than 5 watts PEP.
AD9835 is a simple but well performing 10-bit DDS chip made by Analog Devices (AD). It is controlled via 3 SPI lines transmitting the frequency data. Maximum output frequency is around 16MHz when the chip is clocked with its maximum clock rate of 50 MHz. Oscillator output voltage is some hundred millivolts peak-to-peak, so you can connect the output directly to pin 6 of the NE612 mixer.
Control signals come from an Arduino Pro Mini board. The microcontroller in this module is, if you are an Arduino user, preinstalled with a bootloader program. I overwrote this small portion of code and use the ATMega168, which is the core of the Arduino, in “native” mode. My software is written in C and transferred via “AVR dude” software using the ISP lines MOSI, MISO, SCK and RESET. These lines are not in the schematic, please refor to ATmega168 data sheet. Alternatively you can use, like shown in the schematic, an ATmega168 controller. So you have to de neccessary wiring on your own.
You will find the source code here. I packed it into an Open Document Text File because of problems I encountered when I tried to store the code into this Blogtext. If you need a compiled HEX-file, please feel free to email me!
Display is a very small OLED with 64 by 32 pixels. The OLED is, to my point of view, a little bit noisy. To suppress any rf traveling on VDD line I use an 82 ohm resistor and a set of bypass capacitors of 100uF and 0.1uF capacity closely connected to the OLED VDD pin to GND.
A low pass filter by the output of the DDS ensures spectral purity and avoids clock oscillator feed through. Remember that if you need another output frequency other than 2 MHz you should redesign the low pass filter.
Tuning is done by a rotary encoder connected to PD5 and PD6 of the microcontroller. I use the pull up resistors internal to the microcontroller, so you won’t see any other things than the mere encoder.
Tunings steps are selected by pushing the encoder knob or another suitable push button. This button is connected to ADC0 in the ATMega168 via a 3.9k resistor. The resulting ADC voltage might be problem because of a certain variation in the values of the pull up resistors that form the second resistor of the voltage divider. There is an outcommented section in the code that will show you the exact ADC value that has to be typed into the code so that key recognition works exactly.
The button once pushed will increase the tuning step by a certain amount of Hz. Steps are 10, 50, 100 (standard step), 250, 500, 1000 and 5000 Hz in and endlessly revolving chain. The step will be reset to 100Hz (standard tuning step) by leaving the tuning knob idle for 2 seconds. That’s all with the controls. Very simple, but sufficient.
The transceiver is constructed on a double sided veroboard with 6 by 8 centimeters area. Components are through hole and SMD where available. The Arduino is mounted to the front panel (another Veroboard carrying the controls etc.) as well as the OLED is. The veroboard is inserted into an aluminium frame connected to the front panel with 4 lateral M2 screws:
Wiring can be made by using the colored lines stripped from old parallel printer cables. These cables have a diameter of precisely 1mm an fit through the holes of the veroboard excactly.
If you connect any external components that are not on the same veroboard use standard 2.54 mm (0.1″) male and female board connectors! This will make it much easier to dismantle and reassemble the rig in case troubleshooting is neccessary.
Use M2 srews instaed of M3 when building very small rigs like this one!
The reverse side of the main arrangement:
Two brass made bends (from the local hardware store and each cut to a length of 8 centimeters) hold the PCB inside the mounting frame. A winding has been cut into the brass to fix the bends with screws in M2.
Together with 2 halves of a bent aluminium cabinet covered with “DC-fix” (a German manufacturer of self-adhesive PVC coating) the final rig looks like that:
DK7IH QRP SSB 40m/7MHz pocket size transceiver – final assembly
So, that’s the end of the story so far. Now it’s time for going outdoor and test the rig in field use. 😉
This project tries a new personal approach in designing a very small (i. e. a micro) QRP radio. And also new is the way I want to report about it. The blog entries will be published more or less simultanously to the building progress of the respective modules.
1 A brief project description
The main objective is to set up a SSB QRP transceiver for 6 HF bands (similar to my 5-bander introduced in 2015) now starting with 160m, then 80m, 40m, 20m, 15m and 10m at last.
Another idea I have in mind is to build the rig from separate modules for each single stage so that each main circuit (mainly receiver and transmitter section) is constructed with the needed stages on verobaords that are mechanically identical.
The idea behind that is that a board which does not show top performance in function (or even completely fails) can be changed quickly and an improved version can be installed easily without the need to throw the whole receiver (for example) into the junk box labelled with “failed projects”.
Band switching will be done by small relays again (I purchased 60 SMD relays for 40€ some months ago). The band filter section will be shared by tx and rx section this time. This saves space and reduces effort.
The transmitter will be a 4 stage unit again (more or less the same like in my 5 bander). Output power projected is 5 to 10 watts on all the bands.
The receiver is designed once again as a single conversion superhet because experiments with double conversion were not successful due to a large number of spurs audible in the receiver.
The first mixer is set to be an SBL-3 diode ring mixer. This will give the receiver a very good handling of strong signals, I hope. IF amplifier will be a two staged one with dual gate mosfets controlled by an audio derived AGV voltage. The rest? The usual suspects, I would say. Wait and see!
1.1 The VFO module
I have become quite familiar with SiLab’s Si5351 oscillator module the recent months. I first used it in my “Micro 20-3” trx which was a success. The module is very small, completely ready for use (I’m still using a breakout board made by Adafruit) and able to handle 5V. It provides 3 oscillators that can be programmed independently to put out something lieke a square wave ranging from 8kHz to 160MHz. I have developed a software that avoids any tuning noise, so, this oscillator (which is a clock oscillator by intention) can be used as a VFO for an amateur project.
To keep the effort simple, I reused the 1306 oled module that you can see in lots of my previous projects. Both boards (Si5351 and 1306 oled) are controlled by I²C-bus which allowes me to use a relatively simple micro controller. In this case again I have the Arduino Pro Mini containing an ATMega168 controller (16 MHz) on board. If it should turn out that I might need more memory, the same board here is on stock carrying an ATmega328 controller. Let’s see how this will work out.
This is the circuit of the complete VFO module:
The module will be placed behind the front panel.
Tuning will be done by a Bourns optical encoder that turns very smoothely. I purchased some for under 5 Euros each from pollin.de. An unbeatable price! Unfortunatley they have been quickly sold out.
The core of this module is the Arduino Pro Mini microcontroller centered on the diagram. It is connected to the Si5351 breakout board and the 1306 oled display by I²C bus.
Si5351: Output 0 is used as VFO terminal and output 1 carries the LO signal with 9Mhz. To avoid digital noise spreading on the +5V line a 100µF capacitor should be switched close to the VDD terminal. Proper and short grounding also is recommended to avoid spurs.
OLED1306: Also a 100µF electrolytic capacitor has been added including a 10µH rf choke forming a low pass filter together. I found that these oleds a very prone to distribute digital noise via VVD line, so this measure contributes much to keep your receiver clean from any unwanted signal spektrum generated by the oled.
Keep in mind to tie SCK and SDA lines to +5V via two resistors of 4.7kOhms each!
Band switching: It is software controlled. To save output ports I did not connect the 6 relay drivers for the 6 bands directly to the ports of the microcontroller. I’m using an open collector BCD to DEC driver (74LS145) instead. Ports PB0, PB1 and PB2 are forming a 3-bit pattern that is switched to 6 output lines (output 0 to output 5) of the BCD2DEC driver IC. 74LS145 is capable of handling switch voltages up to 15V thus working with 12V coil relays is easy.
User control interface: This rig has 4 different switches that will be explained later from the functional point of view. The operator can set nearly all functions of the transceiver by using these push buttons and the main tuning wheel. The buttons switch to GND by 4 different resistors and are read by PC0 port of the micro. PC0 equivalents to channel 0 of the integrated analog-to-digital converter (ADC) inside the ATMega168. This also saves controller ports to a large extent (using 3 instead of 6 ports!). So, all in all, I think I can dare controlling a multibander by a relatively small microcontroller.